NOISE FILTER

• Document Type: CREST
• Collection: General CIA Records
• Document Number (FOIA) /ESDN (CREST): CIA-RDP81-00120R000100030008-6
• Release Decision: RIFPUB
• Original Classification: K
• Document Page Count: 15
• Document Creation Date: December 20, 2016
• Document Release Date: February 22, 2001
• Sequence Number: 8
• Publication Date: April 9, 1974
• Content Type: CONT

Approved For Release 2007/09/21: CIA-RDP81-00120R000100030008-6
United States Patent [ 19] [ I I ] 3,803,357
Sacks [45] Apr. 9, 1974
[54]
NOISE FILTER
[76]
Inventor:
Jack Sacks, 815 Tamlei St.,
Thousand Oaks, Calif. 91360
[221
Filed:
June 30, 1971
[21]
Appl. No.: 158,519
[52]
U.S. Cl ................................... 179/1 P, 179/1 D
[ 51 ]
Int. Cl ............................................. H04r 27/00
[581
Field of Search ........... 325/473, 474, 477, 480;
179/1 P
[ 561
References Cited
UNITED STATES PATENTS
3,403,224
9/1968
Schroeder ........................... 179/1 P
3,180,936
4/1965
Schroeder ........................... 325/480
R27,202
10/ 1971
Kahn ................................... 325/477
3,160,707
12/ 1964
Meyers ................................ 179/1 P
3,497,812
2/1970
Dixon ................................. 325/480
a noisy signal content is fed to a plurality of contigu-
ous narrow band nonlinear filters connected in paral-
lel. Each filter has a controllable discrimination
threshold and together cover the audio spectrum
where noise signals are considered objectionable. A
noise tracker connected to the same signal source de-
tects the noise level whenever the desired signal is ei-
ther absent or substantially reduced. The discimina-
tion threshold of each of the narrow band filters is
controlled by the output of the noise tracker, which
thereby controls the ability of each of the narrow band
filters to pass a signal as a function of the noise signal
being detected. The outputs of each of the narrow
band filters are connected together and fed to a com-
bining circuit where the spectral power in the output
of all of said filters is combined in the power phase re-
lationship. The gain of the individual narrow band fil-
ters is reduced in the presence of noise. In the pres-
ence of a strong desired signal the gain is not attenu-
ated and in this manner the signal to noise ratio of the
signal is improved.
Primary Examiner-Kathleen H. Claffy
Assistant Examiner-Douglas W. Olms
[571 ABSTRACT
A composite signal having a desired signal content and
10 Claims, 9 Drawing Figures
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PATENTEDAPH 91974 30803,357
SHEET 1 OF 5
Input
Signal
Pre=
Amp,
Low Pass
Filter
12
16a
V
14
15 I
_ Summing
l Amp.
I 16
Channel (n-2)
# I
/17 I
f-- I
18
Channel (n)
High Pass
Filter
13
Fig. 1.
20
7
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PATENTED APR 91974
SHEET 2 OF 5
Fig. 2.
+ Control
28
L from Logic Circuit
Fig. 3.
Input
from
Pre-amp
Input
NBF
3,803,357
_26
27
Output
NBF
Output
to
Summing
Amp.
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PATENTED APR 91974
SHEET 3 OF 5
Amplitude
Response
Fig. 5.
Channel Channel Channel Channel Channel
1 2 3 n-1 n
Fig. 4.
3,803?357
High Pass Filter
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PATENTED APR 9 1974
3,803,357
SHEET 4 OF 5
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PATENTED APR 91914 3,803,357
SHEET 5 OF 5
Input
Signal
Low Pass
Filter
K/Nc(t)
9~ 96.
Nnrrnw I inAnr
V90
Channel 1. )
Rectifier
6 Shaper
Linear
Multiplier
98
96n
Channel n
95n
Kn
97n 93n
Linear
Multiplier
Narrow
Pass Filter
Linear
Multiplier
Rectifier
a Shaper
J High Pass
Filter
/-98n
80
L
Input
Signal
High
Pass
Filter
Fig. 9.
/ 81
Linear
Multiplier
Full Wave
Rectifier
L
V~
Summing
Amp.
Fig. 8.
J
84
Gate
(normally
closed)
1 185
-~ Integrator
- - - - -J
from "OR" Logic
70 Fig. 7.
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3,803,357
1 2
This invention relates to a process and means for sub-
stantially removing wide band noise contained in the
same audio spectrum as the desired signal.
In the recording art as practiced today, great use is
made of the dubbing procedure where an individual
channel is first recorded and then subsequent channels
are added to the first channel which thereby enhances
the sound and allows the recording engineer and artist
great liberty and flexibility in enhancing the sound. Un-
fortunately, each time a new channel is added to a prior
sound track, broad band noise together with the de-
sired signal is also added to the channel, In many situa-
tions where the noise level is high, the broad band
noise contained in the individual signals can be toler-
ated and is not unduly offensive. However, there are
many situations especially in quiet passages and in soft
renditions where the broad band noise is extremely
harsh and can be heard by the individual listener. Ef-
forts to remove this broad band noise have not been
successful until this invention.
This same problem exists in the movie industry where
the dubbing technique is also used since background
audio information is usually recorded on site and then
placed on the film track at a later time. The action
items are then recorded and at still a later time the ac-
tual voices of the actors and actresses are added to the
already complicated sound track. It must be recognized
that the addition of each new sound track adds with it
a component of broad band noise from that particular
track which generally has the effect of reducing the sig-
nal to noise ratio of the signal.
In the present invention there is described a com-
pletely adaptive system which receives the composite
signal comprising the desired signal and the noisy com-
ponent signal. The basic circuitry comprises a plurality
of contiguous nonlinear narrow band filters, each made
responsive to the amount of noise being detected. In
the presence of a noisy signal, the output will be dimin-
ished whereby in the presence of a strong desired signal
the output is unchanged and the signal will pass undis-
trubed, thereby effectively controlling the signal to
noise ratio of the output signal. It has been recognized
that generally the noisy component is. not contained in
all of the audio spectrum but rather is contained in se-
lected mid-range portions of the audio spectrum. For
example, the low frequency spectrum generally con-
tains a substantially small component of broad band
noise which is not normally considered objectionable.
The main noisy signals are usually contained in the so-
called mid-range and it is here where the majority of
the noisy signals are accounted for and must be re-
moved. Above the mid-range frequencies to the end of
the audio range the noisy signals do not generally
cause a problem.
In the preferred embodiment the signal source is fed
to a plurality of contiguous nonlinear narrow band fil-
ters connected in parallel with each other. Each of the
narrow band filters is arranged to have a controllable
discrimination threshold.
A noise tracker connected to the signal source de-
tects the noise level in the circuit when the desired sig-
nal is either present or very low.
The output of the noise tracker continuously controls
the discrimination threshold and hence the gain of each
of the narrow band filters in response to the noise being
detected. In this manner each of the narrow band filters
is made to discriminate the signal passing through its
filter based upon the presence of noise contained in the
signal source. In other words, in the presence of a noisy
5 signal the discrimination threshold of each of the nar-
row band filters is made larger so as to discriminate
against and prevent the transmission of the noisy signal
by reducing the gain. However in the presence of a
strong output signal, the discrimination circuits have
10 less effect and hence each of the narrow band filters is
free to pass this complete and strong composite signal.
The output of each of the narrow band filters is fed to
a summing amplifier which combines the spectral
power in the output of each of the narrow band nonlin-
15 ear filters.
In the preferred embodiment it will not be necessary
to construct a plurality of narrow band nonlinear filters
from the lowest audio frequency to the highest audio
frequency desired. Experimental evidence indicates
that broad band noise is not a significant problem in the
lower frequencies nor in the higher frequencies due to
psychoacoustical hearing limitations. In the preferred
embodiment therefore a single, low-pass filter covering
the band of spectral frequencies from the lowest fre-
quency to a mid-range frequency where noise is a prob-
lem may be used. The low-pass filter is connected to
the signal source in parallel with the plurality of narrow
band nonlinear filters which cover the mid-range fre-
quencies where noise is a significant problem. A high-
pass filter is connected to the signal source and in par-
allel with the low-pass filter and the plurality of narrow
band nonlinear filters will pass the higher frequencies
where noise is generally not considered a problem. The
output of all of the defined filters is connected to a
summing amplifier where the spectral content in the
output of each of the filters is combined in the proper
phase relationship. The actual cross over points of the
low-pass filter and the high-pass filter will be the func-
tion of the equipment used and the severity of the noise
and, of course, the spectral content of the noisy signals
encountered.
For the worst situation the complete audio band may
be broken up by means of a plurality of contiguous nar-
row band nonlinear filters. However, the process of
combining and controlling the discimination threshold
of each of the narrow band filters would be the same
as mentioned before.
Further objects and advanrages of the present inven-
tion will be made more apparent by referring now to
the drawings which describe the preferred embodiment
and an alternate embodiment. Reference now being
made to the accompanying drawings wherein:
FIG. 1 is a block diagram of the preferred embodi-
ment for this invention;
FIG. 2 is a schematic diagram illustrating a first em-
bodiment of the nonlinear narrow band filter having a
discrimination threshold circuit;
FIG. 3 is a schematic diagram illustrating a second
embodiment of the narrow band filter illustrating a pre-
ferred embodiment for determining and controlling the
threshold of each of the narrow band) filters;
FIG. 4 is a wave form illustrating the action of the
narrow band filter having a controllable threshold por-
tion;
FIG. 5 is a wave form illustrating a composite noise
spectrum showing the effect of the low-pass filter, the
narrow band filters and the high-pass filter and their re-
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3,803,357
3 4
lationship to the desired signal and the broad band sig- In operation the presence of the noisy signal will be
nal; detected by the noise tracker 21 which will generate an
FIG. 6 is an input-output transfer characteristic illus- output signal that will control the discrimination
trating-the output of the narrow band filters in response threshold for each of the individual narrow band non-
to changing conditions of noise and the effect of the 5 linear channels 14 through 18. In the presence of a
noise on the controllable threshold and hence the ulti- noisy signal as detected by the noise tracker 21, the in-
mate effect on the output of the individual filter; dividual threshold will be changed thereby reducing the
FIG. 7 is block diagram of the preferred noise tracker gain of the individual channels. In the presence of a
used in FIG. 1; strong desired signal, the individual channels will be
FIG. 8 is a block diagram of a second embodiment of 10 unaffected and in this way, the signal to noise ratio of
the invention; the complete output signal will be effected and
FIG. 9 illustrates an alternate noise tracker for use changed.
with the circuit illustrated in FIG. 8. Referring now to FIG. 2 there is shown a schematic
Referring now to FIG. 1: there shown a preferred em- diagram illustrating a first embodiment of the nonlinear
bodiment of the present invention. The input signal is 15 narrow band filter illustrated in FIG. 1 as elements 14
generally a composite signal comprising the desired sig- through 18. As mentioned before, each of the channels
nal as well as a noisy component which is generally are identical in circuit form and each channel consists
considered undesirable. This input signal is fed to a pre- of a nonlinear narrow band filter having a discrimina-
amplifier 10 which has the effect of bringing both the tion threshold circuit. A review of FIG. 2 will show that
desired signal and the noisy signal to an acceptable 20 the individual channel is composed of three basic parts,
level for processing. The output of the preamplifier 10 namely an input narrow band filter 25, feeding a non-
feeds a low-pass filter 11, a noise filter 12 and a high- linear threshold device 26, which in turn feeds an out-
pass filter 13 which are all connected in parallel. The put narrow band filter 27. The input to the narrow band
low-pass filter 11 passes the low audio frequencies from filter 25, is from the pre-amplifier 10 illustrated in FIG.
the lowest frequency desired to an intermediate fre- 25 1, whereas the output from the output narrow band fil-
quency. The noise filter 12 comprises a plurality of ter 27, is to the summing amplifier 20, in FIG. 1.
contiguous narrow band nonlinear channels 14, 15, 16, The input narrow band filter 25, and the output nar-
17 and 18 which are all connected in parallel. The row band filter 27 have substantially identical transfer
number of individual channels will be function of the characteristics which approximates that of an interme-
severity and frequency location of the undesirable 30 diate Q (for example, 2 - 6) tuned circuit. The effect
inter-
noisy y signals. As As mentioned pif the signal is of the input narrow band filter 25, is to reduce inter-
nos signl and covers previously,
extremely broad band modulation distortion since the 0 attenuation charac-noisy from the lowest frequency to .the highest audio fre- teristic of the narrow band filter substantially prevents
other desired, then the complete audio system will 35 er signals from passing through the n pass
consist of a plurality of contiguous individual narrow band of the filter. Intermodulation betwee a rrow n the noise
band nonlinear channels. signals and the desired signals will produce cross modu-
lation in the nonlinear threshold device 26. These fre-
For the general application of the present invention, quencies will be outside the band of the filter 25 and
the number of individual channels will cover only the will be strongly attenuated. On the other hand, any sig-
mid-range frequencies where the noise is generally con- 40 nals substantially close to the desired signal which is
sidered excessive and must be controlled. Frequency within the band pass characteristics of the filter 25 will
response of the high-pass filter 13 will cover those produce sum and difference frequencies outside of the
higher frequencies above the highest frequency of band pass characteristics of the filter 27 and hence,
channel 18 and up to the highest audio frequency de- they too will be strongly attenuated.
sired where noise is generally not considered a prob- 45 In the presence of a strong desired signal, any unde-
sired noisy signal passing through the filter 25 with the
The output of all of the filter, namely low-pass filter desired signal will be completely masked. This masking
11, and all of the individual narrow band channels 14, effect takes place in the presence of a substantially
15, 16, 17 and 18, and the high-pass filter 13, are con- strong complex sound signal which as a broad band
nected together and fed to a common summing ampli- 50 noise component. The effect is sometimes called a psy-
fier 20, where the spectral content located in the out- cho-acoustical masking property of the ear in hearing
put of each of the filters is combined. a large complex sound and this invention takes advan-
Also connected to the output of the preamplifier 10, tage of the propensity of the human ear and brain to re-
is a noise tracker 21. The noise tracker 21 generates a duce the effect of any broad band noise component in
signal in response to the noise level whenever the de- 55 the presence of a strong complex sound signal. How-
sired signal is at a minimum or is substantially absent. ever, should the noisy signal be strong and the desired
The output of the noise tracker 21, will therefore be signal weak, and both at substantially the same fre-
a controlled signal that is a direct function of the noise quency so as to be passed by the band pass characteris-
contained in the composite incoming signal. The out- 60 tics of the input narrow band filter 25, then there will
put control signal from the logic circuitry 23, is used to be no masking effect and the wide band noise will come
control the discrimination threshold of each of the indi- through to the nonlinear threshold device 26. It can be
vidual narrow band nonlinear channels 14 through 18. shown that a complex broad band signal will mask out
The output from the noise tracker 23, is actually fed a broad band noisy component and also that a narrow
through a separate weighting network 14a, 15a, 16a, 65 band desired signal will mask out a spectrally similar
17a and 18a associated with each of the individual narrow band noisy component signal. The most adverse
channels 14 through 18, in order to compensate for situation however, is the presence of a narrow band sig-
known variations in the noise spectrum. nal with a broad band noisy component since in this
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situation, there is no masking effect and the broad band
noise will come through.
The controllable nonlinear threshold device 26, has
a band pass characteristic in which the output signal is
controlled by varying the dead zone or the threshold in
response to the presence of noise as determined by the
noise tracker 21 in FIG. 1. In the presence of a noise
signal as detected by the noise tracker, the output of
the nonlinear threshold device 26, is controlled to
widen the dead zone symmetrically about the center
frequency and in this way reduces the gain of the signal
fed to the output narrow band filter 27. The output of
the noise tracker 21 illustrated in FIG. 1, therefore con-
trols the dead zone of the nonlinear threshold device as
a function of the noise content of the incoming signal.
Since the envelope of the incoming signal is symmetri-
cally affected, the overall band pass characteristic of
the complete channel is therefore a function of the
band pass characteristic of the input narrow band filter
25, and the output narrow band filter 27.
For simplicity of explanation, we have assumed that
the band pass characteristics of the narrow band filter
25, and the narrow band filter 27, have been identical.
This is not a requirement since in the preferred em-
bodiment it may be more desired to cascade the band
pass circuits so as to obtain a broader or staggered tune
effect and in this way increase the band pass character-
istics of the individual channel. Where staggered tuning
is desired, it is obvious that the band pass characteris-
tics of the input narrow band filter and the output nar-
row band filter may not be the same. As mentioned pre-
viously, the input narrow band filter 25 attenuates fre-
quencies away from the center frequency. In other
words the probability of widely separate frequencies
inter-modulating and passing through the filter 25 is re-
duced since frequencies away from the center fre-
quency will be attenuated and prevented from passing
through the filter. When considering two frequencies
so very close together that they enter the input narrow
band filter 25, the sum frequencies will be higher than
the band pass characteristics of the output narrow band
filter 27, and hence, will not be passed by the individual
channel in question. Similarly, the difference frequen-
cies will be so low that they in turn will not pass through
the band pass characteristics of the output narrow band
filter 27. Hence, it can be shown that the combination
of the input narrow band filter 25, and the output nar-
row band filter 27, substantially reduce the probability
of intermodulation distortion taking place.
The output narrow band filter 27 performs the func-
tion of cleaning up all irregularities in the signal due to
the presence of the dead zone in the nonlinear transfer
device 26. As mentioned previously the noise tracker
21, in FIG. 1, controls the dead zone variation of the
nonlinear threshold device 26 about the center fre-
quency and in this manner the output signal is symmet-
rical about the center frequency even though some
center portion is removed. It can be shown from a Fou-
rier analysis that any essentially symmetrical signal is
composed mainly of odd harmonics with very few
evens. The lowest odd harmonic having a substantial
energy content is the third harmonic and hence, the
output narrow band filter 27, is designed to have a band
pass characteristic that highly attenuates the third har-
monic, thereby getting rid of any harmonic distortion
that may have been introduced in the nonlinear thresh-
old device 26, by the action of opening the dead zone.
3,803,357
6
Higher odd harmonics are even more severely attenu-
ated.
The width of the dead zone and the total number of
individual channels used in any system becomes a func-
5 tion of the total amount of noise that the designer is
willing to allow to pass through the system. It can be
shown mathematically that the noise power is propor-
tional to band width and that increasing the total num-
ber of channels has the effect of decreasing the band
10 width per channel, and as a result the noise per channel
will also decrease. From a theoretical point of view, it
is possible to decrease the noise per channel to zero as
a limiting factor by increasing the number of channels
to infinity. The important practical consideration how-
15 ever, is that by increasing the number of channels it is
possible to significantly reduce the noise power in
every channel since the band width of each channel is
reduced. It must be remembered however, that the sig-
nal is not affected since the signal will come through
the channel at full amplitude. Hence, by using a plural-
ity of individual channels, the dead zone (and thus the
residual distortion) in each channel can be reduced in
proportion to the number of channels used. Where
noise is a less severe problem, fewer channels may be
required. In a system where noise is a major problem,
more channels must be used. In order to remove as lit-
tle as possible of the desired signal it is important to
keep the gap as small as possible and to this extent, use
the maximum number of channels consistent with eco-
nomic requirements. This invention therefore gives the
circuit designer a wide latitude in adapting the inven-
tion to the economic requirements of the system as a
trade-off against the amount of noise that he can toler-
ate in any given system.
The action of the non linear threshold device 26
therefore is to remove a small slice of the signal at the
zero crossing where it can be shown that most of the
noise signal is located. The circuit can be described as
a zero crossing limiter. The circuit therefore becomes
40 an effective way of eliminating and removing noise
from a signal. It will be recognized, however, that a
strong noisy signal can come through the system if the
strong noisy signal is at or near the frequency of the de-
sired signal. As mentioned previously, the noisy signal
45 will then look like another incoming signal which will
then be masked by the larger desired signal by the psy-
cho-acoustic masking effect.
The nonlinear threshold device 26, in the simplest
sense, comprises a pair of controllable biased diodes in
50 a bridge feeding an operational amplifier. The output
from the narrow band filter 25, is fed through a cou-
pling resistor 28, to the juction of a pair of biases diodes
29 and 30. The diode 29, is connected in series with a
55 resistor 31, and diode 30 is connected in series with a
resistor 32, which resistors are joined together and feed
the input of amplifier 33. A positive bias control signal
is fed through resistor 34 to the junction of diode 29,
and resistor 31. In a similar manner, a negative bias
60 control signal is fed through a resistor 35, to the junc-
tion of diode 30, and resistor 32.
In the presence of any bias voltage the input signal
feeding the bridge circuit must overcome the approxi-
mate 0.7 volt drop in each of the diodes 29 and 30. In
65 other words, the bridge circuit consisting of diodes 29,
30 and resistors 31 and 32 will effectively prevent the
passage of any signal that does not have a swing greater
than 1.4 volt since diodes 29 and 30 cannot conduct
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until the 0.7 volt breakdown point is reached for each
diode. In the absence of a bias signal the bridge circuit
will provide a fixed 1.4 volt dead zone. By using suit-
able positive and negative bias controls feeding the in-
termediate points of the bridge circuit, the actual dead 5
zone can be made smaller or larger depending upon the
sense and the magnitude of the bias currents fed to the
bridge circuit.
It will be obvious to those skilled in the art that a
fixed 1.4 volt dead zone is highly excessive when con- 10
sidering maximum voltage swings of 20 volts peak to
peak for the input driving signal. In addition, it is most
desirable to have the dead zone dynamically controlled
by the output of the noise tracker 21 in FIG. 1, to
thereby make the complete system adaptive to the
amount of noise being detected. A reference to FIG. 4
will more fully illustrate the dynamically controlled
dead zone and the linear symmetrical amplification
achieved by the nonlinear threshold device 26. The sys-
tem described and illustrated in FIG. 2 will allow a dy-
namically controlled dead zone to approach 0.1 or 0.2
volts. However, in order to obtain the full benefits of
the present invention it is necessary that a more precise
control be obtained over the dead zone. The limitation
of 0.1 or 0.2 volt dead zone according to the system il-
lustrated in FIG. 2 is a result of the presently available
diodes 29 and 30. A review again of FIG. 4 will show
that as the dead zone becomes-smaller and smaller the
symmetrical linear portions of the curve 40 and 41, will
approach the zero crossing and begin to appear as a sin-
gle linear curve 42, thereby nullifying the effect of the
invention which requires precise control over the dead
zone.
Referring now to FIG. 3 there is shown a second em-
bodiment of the nonlinear threshold device which over-
comes the inherent disadvantage of the biased diodes
mentioned in connection with FIG. 2. The system de-
scribed and illustrated in FIG. 3 will allow a dynamic
controllable threshold device approaching two or three
milivolts which now provides the greater control
needed to more fully achieve the benefits of the present
invention. FIG. 3 illustrates a complete individual
channel consisting of an input narrow band filter 25,
feeding a new and improved nonlinear threshold de-
vice, which in turn feeds an output narrow band filter
27, as described in connection with FIG. 2. The output
of the narrow band filter 25, feeds resistor 46, which
directs the input signal to a bridge circuit and specifi-
cally to the junction of diodes 47 and 48. Diode 47 is
connected series with resistor 49, and similarly, diode
48 is connected in series with resistor 50. Resistors 49
and 50 are joined together to thereby define the bridge
circuit. The output of the bridge circuit is fed to a resis-
tor 51, which feeds the output narrow band filter 27. A
by-pass coupling capacitor 52 is connected across resis-
tor 49, and similarly, a by-pass decoupling-capacitor 53
is connected across resistor 50.
A positive bias control is fed through a resistor 54, to
8
verting) side of the amplifier. The output of the narrow
band filter 25 is fed through a resistor 58, to the input
of narrow band filter 27.
The parameters of the circuit are chosen so that am-
plifier 56, is operated at high gain and such as 100 (40
d b). The actual amplification of the input signal by am-
plifier 56, will be according to the ratio of resistors 57
and 46. In other words, the gain achieved by amplifier
56, will be equal to minus R 57 over R 46. The amount
of signal fed to the input of filter 27 is controlled by se-
lecting the ratio of resistors 51 and 58 to be the same
as the ratio of resistors 57 to 46. In other words, the
ratio of resistors 51 to 58 is the same as resistors 57 to
resistor 46. The DC current in the bridge circuit is bal-
anced by making resistor 54 and resistor 55 in the bias
control system equal. Similarly, resistor 49 and resistor
50 in the bridge circuit are made substantially equal. In
the preferred embodiment the ratio of resistor 57 to re-
sistor 46 was selected to give a gain of approximately
100, or in other words, resistor 57 was 100 times the
resistance of resistor 46, which results in a gain of 100.
Since the ratio of resistors 51 to $8 was made the same
as the ratio of resistors 57 to resistor 46 it follows there-
fore that the resistance of resistor 51 is 100 times the
resistance of resistor 58.
If we consider a signal from input narrow band filter
25, to be ei than with the parameters chosen, the out-
put signal from amplifier 56 at the junction of resistors
49 and 50 will be -100 ei. The signal feeding the filter
27 will consist of the output from ei the bridge circuit
and the ratio of the input signal ei fed through resistors
58 and 51.
The first signal will be R58/(R51 + R58) -100 ei +
R51/(R51 + R58). Remembering that the original con-
dition of the circuit was set up with the ratio of resistors
57 to 46 being equal to 100 and further that the ratio
of resistors 51 to 58 was made the same as the ratio of
resistors 57 to 46 we can show that the circuit will bal-
ance and the output signal will be zero.
With the circuit balanced as shown we have now
proved that for low level signals below the slipping level
that there will be zero output from the nonlinear
threshold device 45. In other words, for noise signals
below the level at which the diodes 47 and 48 operate,
there will be no output from the circuit. This means
that we now have available a low level threshold con-
trol of approximately one one-hundredth of the signal
necessary to cause the diodes 47 and 48 to conduct.
The benefits achieved by the system illustrated in FIG.
3 can now be more fully appreciated over the system
described in connection with FIG. 2. The system of
FIG. 2 was limited to the voltages at which the diodes
would conduct and according to the present date tech-
nology these diodes can be controlled by suitable bias-
ing control to conduct to within tenths of a volt. By
using the circuit described in FIG. 3 it is now possible
to get selective biasing control to within one one-
hundredth of the voltages at which the diodes will con-
duct.
Referring now to FIG. 5 there shown a curve illustrat-
ing a typical noise spectrum covering the low level sig-
nals where noise is generally not a problem. Noise is
considered a problem in the mid channels whereas
noise is less audible at the higher and lower frequen-
cies. The band pass frequencies of the low-pass filter,
the individual channels comprising the noise filter, and
the high-pass filter are more graphically illustrated to
the junction of diode 47, and resistor 49, and similarly 60
a negative bias control is fed through a resistor 55, to
the junction of diode 48, and resistor 50. An integrated
circuit/operational amplifier 56, (op-amp), has the
negative gain achieved input connected to the junction
of diodes 47, 48 and resistor 46. The output of the am- 65
plifier 56, is connected to the junction of resistors 49,
50 and 51. A feed back resistor 57, is connected from
the output of the amplifier 56, to the negative input (in-
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show the relationship between all filters covering the
complete audio spectrum.
Referring now to FIG. 6 there is shown a graph illus-
trating the amplitude characteristic of an individual
noise filter channel at a point after the output of the
second narrow band filter. The curve shows that in the
presence of a strong input signal (of high amplitude) a
very small slice, will be removed from the input signal
and hence, nearly all of the signal amplitude in the indi-
vidual channel will be available. Curve 60 shows a high
input amplitude with very little attenuation of the indi-
vidual channel. In the presence of a low input ampli-
tude signal which has a noisy component the operation
of the individual channel will be to limit the amplitude
of the signal passing through that particular channel
since a larger dead zone will be present and hence, less
amplification of the signal will be available. This effect
is shown by Curve 61 which illustrates a low input am-
plitude signal. The effect of the noise filter is very simi-
lar to that of an expander and compressor circuit with
the advantage however, that a controlling DC signal is
not necessary since the level of the input signal itself
dynamically controls the gain of the individual channel.
Referring now to FIG. 7 there is shown a preferred
embodiment of the noise tracker illustrated in connec-
tion with FIG. 1. In order to appreciate the significance
of how the noise tracker operates, it is best at this time
to consider a composite signal containing a desired sig-
nal and a noisy component. A review of the spectral
content of most musical instruments will show a sub-
stantially strong fundamental wave plus even and odd
harmonics that attenuate as the frequency increases.
This is generally true except as regards some percussive
instruments. Since the main power of most desired sig-
nals is in the fundamental frequency we can measure
the intervals between the zero crossings to detect a pre-
dominance of low frequency components of the signal
since there is generally more power or amplitude in the
lower frequency components than the high frequency
components. On the other hand analysis of the zero
crossings of a noisy signal will show zero crossings ran-
domly distributed over the frequency range without a
falling off as frequency increases as is detected in a mu-
sical signal.
The zero crossings of noise will be statistically closer
together indicating a generally higher order of frequen-
cies. Remembering that the fundamentals of most mu-
sical instruments is relatively low in frequencies and
generally below 5,000 Hz., we can now appreciate that
musical instruments will therefore have a less random
and statistically wider spacing between zero crossings
as opposed to the noise signals. Observation of the
spectral content of musical instruments have confirmed
that the spectral content of musical instruments does
generally roll off at the higher frequencies while noise
signals remain generally flat and sometimes increase.
These observations and a statistical analysis have con-
firmed the fact that the zero crossings on the average
from musical instruments are therefore further apart
then zero crossings associated with noise.
A noise tracker, therefore is arranged to generate a
signal in proportion to the time between zero crossings
as a means of measuring and differentiating a desired
musical instrument signal from a noisy signal. The noise
tracker illustrated in FIG. 7 feeds the input signal
through a high-pass filter to a sample and hold circuit
3,803,357
10
which continuously samples the noise signal. The out-
put of the sample and hold circuit is fed to the individ-
ual channels for adjusting the threshold or dead zone
of the individual nonlinear circuits. In the presence of
5 a desired musical signal the input to the sample and
hold circuit is interrupted and the output held in mem-
ory while the noise tracker identifies the incoming sig-
nal as desired signal. This hold may last as long as 15
to 30 minutes for long sustained musical passages.
10 The input to the noise tracker is fed to a first channel
which has the function of detecting the presence of a
desired signal such as a musical instrument. The first
channel comprises a high-pass filter 65, having a cutoff
frequency starting at approximately 1,000 cycles in
15 view of the previously discussed reason that audible
noise signals will generally appear above the fundamen-
tal frequency when dealing with musical instruments.
The output of the high-pass filter 65, feeds both a zero
crossing detector 66, and full wave rectifier 66a, and a
normally closed gate 67, which in turn feeds a sample
and hold circuit 68. In the normal case, the incoming
signal will be identified as noise and will pass the high-
pass filter 65 then be rectified in 66a, pass through the
normally closed gate 67, and feed the sample and hold
circuit 68, which in turn will operate to adjust the
threshold gate of the nonlinear detectors comprising
each of the individual narrow band channels. The sys-
tem being described will identify the desired signal as
either being music or desired sibilant, which will have
the effect of opening the normally closed gate 67,
thereby interrupting the input reading upon the sample
and hold circuit 68. In this manner sample and hold cir-
cuit 68, will control the threshold by memory until the
next reading as determined by the control on the gate
67.
The zero crossing detector 66, in the present applica-
tion functions as a hard limiter since it has an extremely
high gain but a small dynamic range. In this mode it is
possible to obtain a desired output at the time of zero
crossing even in the presence of high amplitude signals.
The output of the zero crossing detector 66, will actu-
ally be a square wave having a repetition rate depend-
ing upon the rate of zero crossings detected. The out-
put of the zero crossing detector is fed to a differenti-
ator and a full wave rectifier 67a, which produces a plu-
rality of positive going spikes corresponding to the lim-
ited or changing square wave generated by the zero
crossing detector 66. The output of the differentiator
and full wave rectifier 67a, is fed to an integrator and
50 filter 67a, that generates a DC voltage having an ampli-
tude depending upon the frequency of the individual
spikes feeding the integrator and filter circuit 68a.
In circuits of this type the individual spikes will cause
a capacitor (which forms part of the integrator and fil-
ter ter circuit 68a) to discharge and in this manner the ra-
pidity of the spikes from the differentiator and full
wave rectifier circuit 67a, will directly affect the magni-
tude of the DC signal coming from the integrator and
60 filter circuit 68a. The DC signal output from the inte-
grator and filter 68a, is smoothed and filtered and now
represents in magnitude a function of the spacings of
the individual zero crossings as detected by the zero
crossing detector 66. In other words, the amplitude of
65 the DC signal will be inversely proportional to the spac-
ings of the detected zero crossings. The DC signal is fed
to a threshold comparator circuit 69, (which is actually
an amplitude comparator) which in effect compares
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the input DC signal against a fixed reference DC signal.
In the presence of a musical signal input the output of
the threshold comparator 69, is a function of the level
of the amplitude of the fixed reference signal. The level
is chosen so that in the presence of a musical signal
input an output signal from the threshold comparator
69, will be fed to an OR logic circuit 70, which will
open normally closed gate 67, thereby preventing the
sample and hold circuit 68, from identifying the signal
as being noise.
The circuit just described therefore has the capability
of specifically identifying the presence of a musical sig-
nal and opening a gate 67, in the presence of this de-
tected musical signal.
As mentioned previously, there are sibilants andd
other signals that look like noise but are in fact desir-
able signals in the voice range that should be identified
as desired signals and should not be discriminated as
noise. The second channel of the noise limiter identifies
and processes these sibilant sounds.
Since the sibilant signals are statistically and spec-
trally similar to the noisy and undesirable signals, it is
not possible to discriminate against these sounds by
;means of the zero crossing technique mentioned above
for the first channel. It is known, however, that sibilant
information does come through as part of the compos-
ite signal as a rapid increase in amplitude or a burst of
signal. In addition, this information is also at a higher
frequency usually above 5,000 or 6,000 Hz. The sec-
ond channel is therefore connected to the same input
as before and comprises a high-pass filter 71, which is
preferably arranged to pass frequencies above 5,000
cycles. The output of the high-pass filter 71, is first rec-
tified by rectifier 72, and then averaged by means of a
low-pass filter 73. If the signal is basically noise it will
be statistically constant and the output of the rectifier
72, will therefore be an essentially constant rectified
signal. The low-pass filter 73, will smooth the signal and
generate a substantially constant DC signal which will
have an amplitude representative of the level of the
rectified input signal. The time lag of the low-pass filter
73, will be substantially long of the order of a tenth of
a second. The output of the low-pass filter 73, is fed to
a scaling network 74, which for example will amplify
the DC signal by a factor of 2. The output of the scaling
network is fed to an amplitude comparator 75, which
receives a second signal directly from the output of the
rectifier 72.
In operation the output of the scaling network will
continuously compare the output of the rectifier 72, so
that in the presence of a sibilant or cymbol crash or a
large burst of amplitude will be detected by the ampli-
tude comparator as an immediate change between the
two inputs. It is true that over a period of time the out-
put of the low-pass filter 73, will rise and approach the
output of the rectified signal from rectifier 72. How-
ever, because of the differential time lag between the
two signals the difference will be immediately detected
at thee output of the amplitude comparator 75. The
output of the amplitude comparator 75, is fed to a sin-
gle shot multi-vibrator 76, which immediately gener-
ates a signal that is fed to the OR logic gate 70.
The effect therefore is that the presence of a speech
sibilant or similar sound is detected by an increase in
amplitude and an output signal will be generated from
the amplitude comparator 75, which will fire a single
shot multi-vibrator 76, that will generate an output sig-
3,803,357
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nal fed to the OR logic gate 70, which will open gate
7 and again prevent the sample and hold circuit 68,
from identifying the signal as noise.
The noise tracker defined and illustrated in FIG. 7
5 therefore has the capability of identifying and measur-
ing desired musical or voice sibilant signals and identi-
fying these signals as desired signals. In the presence of
a desired signal output the sample and hold circuit 68,
will continuously sample the incoming signal as noise.
10 The noise tracker system may be thought of as a fail
safe system since the desired signal is positively tracked
and identified. However, in the event that a noisy burst
is identified as a desired signal, the only effect is that
the gate 67 is opener) and the signal is identified as a de-
15 sired signal and hence, the signal is not lost but rather
is passed through the system. The noise tracker moni-
tors the noise content of the input signal and adapts the
threshold or gain of the channels in response to the de-
tected noise. A review of the embodiment of the noise
tracker described in connection with FIG. 7 will show
that the output of the sample and hold circuit 68, is a
signal that is directly proportional to the measured
noise. Therefore, in the presence of a noisy signal the
output from the sample and hold circuit 68, will be
greater and hence, a larger signal will be required to
"open up" the individual nonlinear circuits comprising
the individual channels as illustrated in connection with
FIG. 1.
The second embodiment is more fully illustrated in
30 FIGS. 8 and 9 and operates in a feed forward mode
very similar to an automatic gain control circuit. A re-
view of FIG. 7 will show that the output of the sample
and hold circuit 68, will be directly proportional to the
spectral content of the detected noise signal. Should
35 the sample and hold circuit 68, detect a large level of
spectral noise, then an increased signal will be gener-
ated which signal will directly increase the threshold
dead zone of the associated nonlinear filters. In other
words, an increased level of detected noise signal will
40 mean an increased control over the associated nonlin-
ear filters.
In the system to be described in connection with
FIGS. 8 and 9, a reciprocal noise signal is generated
which signal is fed back to the input of the individual
45 narrow band circuits so as to reduce the input signal
gain in proportion to noise in the presence of an incom-
ing signal. Referring now to FIG. 9 there shown a block
diagram of a noise tracker which utilizes many of the
50 circuits illustrated in connection with FIG. 7 to gener-
ate a signal representative of the reciprocal of the noise
signal and referred to as K/Nc(t). The input composite
signal contains the desired component Sc(t) and the
noisy component N(t) and is fed to a high-pass filter
80, which has a low frequency cutoff of approximately
55 1,000 Hz. That portion of the composite signal above
1,000 Hz. will pass the high-pass filter 80, and be fed
directly into one terminal of a linear multiplier 81. The
output of the linear multiplier 81, is fed to a full wave
60 rectifier 82, which generates a DC envelope signal
which follows the amplitude of the incoming signal. A
reference signal from source 83 is combined with the
DC output from the full wave rectifier 82, to produce
a difference signal which is fed through a normally
65 closed fast acting gate 84. Gate 84 is controlled by
identical circuitry to that illustrated in FIG. 7 which is
used to control fast acting gate 67. The operation or
control of the gate 84, is such that gate 84 will be held
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open only in the presence of a desired signal or in the
presence of amplitude detected sibilant signals as de-
scribed in connection with FIG. 7. In other words gate
84 will remain closed in the presence of a noise signal
and open in the presence of a desired signal. The out-
put of the gate 84 is fed to an integrator 85, of the type
that will maintain a charge on the output due to the ac-
tion of the high gain amplifier comprising the integra-
tor. The output of the integrator 85, is fed back to the
linear multiplier 81, and in that way provides the de-
sired reciprocal noise signal of K/Nc(t).
The operation of the circuit described in connection
with FIG. 9 is more fully understood by considering the
following parameters where a desired signal is not de-
tected and hence, there's no output from the OR logic
70 from FIG. 7 to open the gate 84. This condition by
definition means that only a noisy signal is coming
through and hence, the input signal fed to the high-pass
filter 80, will only contain noise previously identified as
Nc(t). The varying noisy signal is fed to the linear mul-
tiplier 81, the output of which is rectified to a DC signal
by the fujl wave rectifier 82. The output DC signal is
differenced from a reference source 83, which differ-
ence signal is a varying DC signal which very closely
follows the instantaneous variations of the incoming
noisy signal. Since the fast acting gate 84 is closed in
the presence of a noisy signal, a difference signal repre-
senting the difference between the instantaneous DC
signal generated by the full wave rectifier 82, and the
reference signal 83, will be fed through the fast acting
gate 84, as an error signal or difference signal to the in-
tegrator 85. The integrator will of course integrate the
error signal and feed the output integrated signal back
to the linearmultiplier 81, in the proper phase so as to
attempt to reduce the error signal generated by the dif-
ference between the full wave rectifier 82, and the ref-
erence signal 83, to zero. A review of the mathematics
will show with the input signal to the linear multiplier
81, being substantially the noisy signal of N(t) that any
feed back signal generated by the integrator which will
null out the error signal generated by the difference be-
tween the full wave rectifier 82, and the reference sig-
nal 83, must be therefore the reciprocal of the input
noise signal or in other words, the feed back signal can
be shown mathematically to be K/Nc(t). The circuit
just described in connection with FIG. 9 is part of the
noise tracker used in connection with FIG. 8 and is
used primarily to generate a reciprocal of the noise sig-
nal which is K/Nc(t ).
If during the operation of the circuit a desired com-
ponent of the signal is detected, the fast acting gate 84
will be energized and opened and as a result the inte-
grator 85, will then hold the last level of input voltage
before the gate 84 opened the input circuit to the inte-
grator. The memory of the integrator 85, will maintain
this signal for a period of time until the next noisy pas-
sage as indicated by the closing of the gate 84 at which
time the output of the integrator again tracks and at-
tempts to reduce the input to zero by generating the re-
ciprocal of the noisy component signal as described.
The system illustrated in connection with FIG. 7 uti-
lizes the reciprocal component of the noise for reduc-
ing the gain of the individual narrow band channels and
in this manner acts as an automatic gain control since
in the presence of a feed back signal of the reciprocal
of the noise component, a greater desired signal is re-
3,803,357
14
quired to obtain the same gain output of the channels.
Referring now to FIG. 8 there is shown a second em-
bodiment of the invention which utilizes a low-pass fil-
5 ter 90, and a high-pass filter 91, which are connected
in parallel to the input composite signal consisting of a
desired portion S(t) plus a noisy portion N(t). The out-
puts of the low-pass filter 90, and the high-pass filter
91, is fed to a summing amplifier 92, for the same rea-
10 sons described in connection with the first embodi-
ment. Considering for example channel 1 for a system
having n channels, the input signal is fed to a narrow
band filter 95, which is tuned to a first frequency and
has a band pass characteristic approximating that of a
15 tuned circuit. The frequency response is very similar to
that as described in connection with the first embodi-
ment and as illustrated in FIG. 5. The output of the nar-
row band filter 95, is fed to a linear multiplier 96, how-
ever, a portion of the output signal from the narrow
20 band filter is fed to a weighting network 97, then to a
linear multiplier 93, and then to a rectifier and shaper
98, which has a fast attack time so that the generated
output signal is a DC signal capable of following the en-
velope variations of the wave form passed by the nar-
25 row band filter 95. The DC signal from the rectifier and
shaper 98, is fed to the linear multiplier 96, with the ef-
fect that in the presence of a large input signal, there
is produced a high amplitude DC signal from the recti-
fier and shaper 98, which tends to increase the gain of
30 the linear multiplier 96, to a maximum gain of unity as
shown in connection with FIG. 6 and specifically in
Curve 60. The linear multiplier 93, also receives an
input of the reciprocal of the noise signal generated
from the output of the integrator 85, in FIG. 9. In other
35 words a first input to the linear multiplier 93, will be a
composite desired and noisy signal whereas the second
input to the linear multiplier will be a. DC signal repre-
senting the reciprocal of the detected noise signal. The
effect of multiplying the DC signal with the composite
40 signal would be to scale the output of the linear multi-
plier by a factor determined only by the reciprocal of
the noise signal. The desired result will be that in the
presence of a high level noise, the reciprocal noise sig-
nal from integrator 85, will he low and hence, the gain
45 of the individual channels will be low.
It must be remembered that simultaneously with this
noise signal from integrator 85 of FIG. 7 will be a large
noise composite signal indicated by a large N(t) passing
through the linear multiplier 93 from the narrow pass
50
filter 95.
The weighting network 97, is included to compensate
for known variations and acoustical unbalances that
can be predicted in advance for each of the individual
channels, and for non-uniform noise spectral distribu-
tions. tions. The over all effect is that in the presence of a
large signal being passed through the narrow band filter
95, there is produced an increased gain from the linear
multiplier 96. If thhe increased amplitude of signal is a
60 desired signal namely S(t), then correspondingly, the
noise will be small and hence, the reciprocal of the
noise signal from integrator 85, in FIG. 7 which is fed
to the linear multiplier 93, in FIG. 8, will be high,
thereby increasing the gain of the linear multiplier 93.
Similarly, the increased signal passed by the narrow
65 band filter 95, will generate a large DC signal drom rec-
tifier and shaper 98, which also increases the gain of
the linear multiplier 96, which is the desired result.
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However, if we now consider the presence of a large
noisy signal which has an increased gain, then from our
prior discussions, we know that the reciprocal of the
noise signal, namely K/Nc(t) from the integrator 85, in
FIG. 7, will be low and hence, the gain of the linear
multiplier 93, will be decreased as shown by Curve 61
in FIG. 6, which represents a substantially low input
amplitude and hence, a low gain output. The effect
being that the linear multiplier 93, now has a reduced
gain inthe presence of a noisy signal. Since the over all
amplitude of the signal has been decreased the DC sig-
nal generated by the rectifier and shaper 98, will be low
and hence, the output of the linear multiplier 96, will
also be low. The over all result is that in the presence
of a noisy signal the gain of the system for low level sig-
nals has been automatically decreased, which is again,
the desired result.
The over all effect of the embodiment illustrated in
FIG. 8 is exactly the same as that shown in connection
with FIG. 1. However, the implementation is different.
In discussing noise values in the specifications it must
be remembered that we are now dealing with noisy sig-
nals that are at least 30 to 40 DB below the maximum
signal. The desired signal will therefore always be much
larger than the noisy signal even in a noisy recording.
The individual channels are duplicated n times for
the n channels that are needed to complete the over all
system. The exact number of channels will of course
depend upon the severity of the noise problem and the
specific bands where the noise predominates. It is envi-
sioned that for a very severe noisy system that the com-
plete band pass may be covered by a plurality of indi-
vidual channels as just described in connection with
channel 1. For the conventional system it is envisioned
that a low-pass filter 90, a plurality of individual chan-
nels and a high-pass filter 91, will be sufficient. The
output of all of the defined low-pass filter narrow band
channels and high-pass filter will be fed to a summing
amplifier 92, which will combine the spectral outputs
in the outputs of each of the defined filters. A review
of FIG. 8 will show that there is no need for a second
narrow band filter in any of the channels as there was
in connection with the first embodiment illustrated in
FIG. 1. The reason for this elimination is the absence
of a nonlinear element in any of the individualchannels
as there was in connection with the system illustrated
3,803,357
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Placing the limiter between the narrow band filters and
in series with the nonlinear element is advantageous be-
cause the signal will be limited symmetrically about the
center frequency and hence, the second narrow band
5 filter which has its attenuation point of the third har-
monic way down on the slope of the band pass charac-
teristic curve will therefore pass a symmetrical or pure
sign wave which is actually the fundamental frequency
since the input signal will be clipped symmetrically
10 about the center frequency and hence, the distorted
component will lie primarily in the amplitude of the
third harmonic which the second narrow band filter
will substantially suppress.
What is claimed is:
15 1. In combination,
a plurality of contiguous non-linear frequency selec-
tive narrow band channels connected in parallel to
asignal source, each channel having a continuously
controllable amplitude threshold,
20 a noise tracker circuit connected to said signal source
for generating a control signal in response to the
noise level when a desired signal is substantially ab-
sent,
means for continuously controlling the amplitude
25
threshold of each of said narrow band channels
with said control signal whereby the ability of each
narrow band channel to pass a signal is continu-
ously controlled, and
means for combining the spectral output of each of
said channels in the proper phase relationship.
2. A combination according to claim 1 which in-
cludes means for individually weighting the amplitude
threshold of each non-linear channel to thereby com-
pensate for known variations in the noise spectrum.
35 3. A combination according to claim 1 in which each
channel comprises a series of circuit having an input
narrow band filter,
an amplitude controllable threshold non-linear de-
vice fed by said input narrow band filter, and
40 an output narrow band filter.
4. A combination according to claim 3 in which the
band pass characteristics of said input narrow band fil-
ter and the band pass characteristics of said output
45 narrow band filter are substantially equal.
5. A combination according to claim 3 in which said
input narrow band filter has a given band pass charac-
teristic and said output narrow band filter has a given
bodiment the second narrow band filter had a band
pass characteristic that highly attenuated the third har-
monic and thereby preserved the fundamental fre-
quency as it passed through each of the individual
channels. In the second embodiment as illustrated in
FIG. 8 the nonlinear element in the individual channels
has been eliminated and hence, there is no need for the
second narrow band filter. The input-output character-
istic of the linear multiplier 96, is always a straight line
even though the DC signal feeding the multiplier will
vary the slope and hence, the gain of the multiplier, but
at all times the linear multiplier 96, will be linear. This
fact is more properly illustrated in connection with the
graph shown in FIG. 6.
Many modifications of the present invention will sug-
gest themselves to those skilled in the art. For example,
in the first embodiment, it may be very desirable to
limit the signal in the individual channels by including
a peak limiter between the two narrow band filters.
50 and all higher order harmonics of the fundamental fre-
quency passing through the first filter are attenuated by
the second filter.
6. A combination according to- claim 1 in which the
outputs of each of the channels is combined by sum-
55 ming the output of each of the channels.
7. A combination according to claim 3 in which the
non-linear device includes a bias controllable diode
bridge circuit for removing those signals identified by
the noise tracker as noise signals.
8. A low noise audio system comprising,
a low pass filter connected to a signal source and
adapted to pass a band of audio frequencies in the
frequency spectrum where noise signals are not
considered objectionable,
a plurality of contiguous non-linear narrow band
channels connected in parallel to said signal
source, each channel having- a continuously con-
trollable amplitude threshold,
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a noise tracker connected to said signal source for
generating a control signal in response to the noise
level when the desired signal is substantially absent,
means for continuously controlling the amplitude
threshold of each of said narrow band channels
with said control signal whereby the ability of each
narrow band channel to pass a signal is continu-
ously controlled
a high pass filter connected to said signal source and
adapted to pass a band of audio frequencies in the
frequency spectrum where noise signals are not
considered objectionable, and
means for combining the spectral output of each of
said channels and said filters in the proper phase
relationship.
9. In a system having a niose tracker and a plurality
of individual frequency responsive channels the
18
method of controlling the output of a channel in the
presence of noise that comprises the steps of;
first detecting and measuring the quantity of noise in
a selected frequency spectrum during the absence
of a desired signal, and
then using the measured value of noise to control the
amplitude threshold at which a signal is passed
through each of the individual channels.
10. In a system having a noise tracker and a plurality
of individual frequency responsive channels a system
for improving the signal to noise ratio comprising;
means for detecting and measuring the quantity of
noise in a selected frequency spectrum, and
means responsive to the measured value of said noise
for controlling the amplitude threshold at which a
signal is passed through each of the individual
channels.
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