JPRS ID: 8859 USSR REPORT ELECTRONICS AND ELECTRICAL ENGINEERING

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ELt~ , ~ _ , ,

10 JRNURRY 1980 CFOUO 1r80) 1 aF 3
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JPRS L/8859
10 January 1980
USSR R~ ~rt
p
- ELECTRONICS AND ELECTRICAL ENGINEERING
_ cFOUO lrso)
~
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JPRS L/8859
10 January 1980
USSR REPORT
ELECTRONICS AND ELECTRICAL ENGINE~RING
_ (FOUO 1/80)
This serial publication contains articles, abstracts of articles and news
iterns from USSR scientific and technical journals on the specific subjects
refl.ected in the table of c~ntents.
Photoduplications of foreign-language sources may be obtained from the
Photoduplication Ser~~ice, Library or Congress, Washington, D. C. 20540.
Requests shouZd provide adequate identification both as to the source and
the individual ar~~c?~(s) desired. -
CONTEHTS PAGE _
ANTENNAS 1
Cylindrical Antenna Arrays with Coherent Optical _
Signal Processing 1
CERTAIN ASPECTS OF TELEVISION, PHOTOGRAPHY AND MOTION PICTURES 9
- Electronic Synthesis of TV Pi~tures 9
COMMUNICATIONS, COMMUNICATIONS EQUiPMENT; NETWORKS; DATA
TRANSMISSION AND PROCESSING 12 =
Waveguide Transmission Lines 1~ .
Average Recurrence Frequency of Noise Overshooting in a E
Digital Reversible Memory 15
~.xperimental Study of Space-Two-Channel Detection Receiver--
Interf erence Compensator 18
- Optimum Reception of Complex Signals with Undetermined _
Modulation Characteri~tics 23
Optimal Reception of Discrete Signals Against the Back- -
ground of Random Pulsed Noise 32
An Optimum Two-Stage.Procedure for Detecting a Signal in Noise . 40 .
Theory of Signal Restoration 45
A Study on the Transnission of ar~ Optical Signal With Mu.lti-
position Pulse-Time Modulation over a Communication Line with -
Repeaters 47
- a- [III - U5SR - 21E S&T FOUO]
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. L~~JI~ V1111.r1a~L VJU V~~u~
CONTENTS (Continued) Page
Optimization of a Procedure for Sequential Detection of the
Delay in a Received Signal 57
Ground and Space Microwave Link Radio Wave Propagation 69
Radio Relay Communications 74
Interference of Side Lobes of a Linear-Frequency Signal
from a Near Target 78
On the Effectiveness of Circular Polarization Antennas in
Systems of Radio Communication with Moving Objects 83
Actual Noise-Immunity of Automatic Selection Devices in -
- Mobile Radio Communications Systems 90
Electrocommunications Equipment and Its Production 97
CONVERTERS, INVERTERS, TRANSDUCERS 101
Electronic-Optical Converters and Their Use in Scientific
Research 101
Electrical Conductivity Sensors 105
ELECTRICAL ENGINEERING ~QUIPMENT AND MACHINERY 108
Mathematical Simulation of External Electromagneric Fields ~
of Sources 108
ELECTROMAGNETIC WAVE PROPAGATION; IONOSPHERE, TROPOSPHERE;
ELECTRODYNAMICS 114
On the Electromagnetic Fiel.d in the Vi~inity of the Edge of a
Conducting Half-Plane 114
Measurement of VLF Signals Reflected from the Ionosphere 119
ELECTRON ANA ION DEVICES 128
,Magnetic Periodic Quadrupole Focusing of Intensive
Electron Beams 128
ELECTRON TUBES; ELECTROVACUiJM TECHNOLOGY 139
Analysis of Nonlinear Signal Distortions in Traveling-Wawe
Tubes 139
GENERAL CIRCUIT THEORY AND INFORMATION 149
Detei-mination of Kinds of Combinat ion Interf erence with a
Polyharmonic Effect on a Nonlinear Device 149
MICROELECTRONIC$ 154
Chemical Industry Equipment in Microelectronics Manufacture 154
- b -
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. Antennas
UDC 621.396.677.8.001.5
CYLINDRICAL ANTENNA ARRAYS WITH COHERENT OPTICAL SIGNAL PROCESSING
Kiev IZVEST'ZYA VYSSHIKH UCHEBNYKH ZAUEDENIY RADIOELEKTRUNIRA in Russiau `
Vol 22 No 5, May 79 pp 29-34 -
[Article A.Yu. Grinev and Ye.N. Voronin, manuscript received 30 Jun 78J
[Text] The specific features of the shaping of the
- receive beams of aplanar antenna arrays by radio-optical
methods which allow for a para11e1 view of space in a
broad sector of angles are treated. The possibility of
the reduction of the coherent optical processor of a
cylindrical antenna array is demonstrated and an example =
is adduced.
- The shaping of the receive beams of aplanar antennas by coherent optical
(KO) methods in an approximation of a continuous aperture was analyzed in
[1]. The results of [1] are extended to discrete apertures in the form of
- antenna arrays [AR's] consisting of rad.iators arranged on an aplanar geo-
metric or conducting surface. _
The "radio value:'image, as follows from [1], is generated in the form of a
continuous fan ~f directional, patterns (DN's) of the antenna F(K, K'),
modulated by the angular spectrum of received radio waves:
E~K~~ - J J~(~ F(K, K') d~, (1)
4Jt
when the aperture response of the antenna is acted upon by the following
operator: � - � - .
_ L . f f . . . ~ ~ ~
= opc (R, K') dzR, ~2~
E
~
where Jopt (R, K') is the optimal (for example, in terms of the side lobe
suppression, directional gain, etc.) amplitude-phase distribution (~,FR) of
the excitation of the aperture for the case of pencil-beam recep*_ion from -
the direction K' (}K' is a wave vector); E is the aperture surface; 1~ is the
current point of the aperture.
1
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Ir_ the case of an anten*-~ array of N radia;:ors, positioned at the points
~n, the directional patte.rn in (1) ??as the form:
N ~3~
FAP ~K, K') F3 (K, Rn) exp - iKR,,.lop? (R,,. ~i'),
n=1
~
w'~ere F3(~, ~)exp -i~n is the directional pattern of the n-th element,
_ ceferenced to the origi*-. of the coordinates.
On analogy with [1], by using (1) and (3) instead of (2), we obtain:
L~ . ~ . . .Toar (RR, K j. (4 )
We will note that similarly to algorithm (2), operator (4) has the equivalent
alternative:
^ ~ ?
~.AP . _ ~ . . . JaDt ~Rn+ K'~ ~5 ~
n~f
, (where the symbol * signifies the complex con~ugate), which in contrast to
(4), generates a focused image of the type (1) in the form of a first order
diffraction negative jl].
It is expedient t~ express opera~or !4) in terms of operator (2) by making
use of the Dirac function d(~, Rn):
N
Lpp . = L . . ~ 8 (R, R,?) (6)
-
As we see, the requisite c~herent optical processing using algorithm (4)
reduces to processing with algorittun {2), carried out by means the cor-
responding KOP [coherent optical processor] for a continuous aperture j7,]~
if in this case the input signal cf the KOP is "weighted" wtth the "Dirac �
comb" ~,b(R,Rn)� � It is obvious that an N-channel spatial-time light
L
n=1
modulator~(PVMS) can serve as the latter, where the modulator channels are
arranged in accordance with the law governing the addressing of the original
- KOP.
It is appropriate to note that algorithm (6) encompasses all the special -
cases of antenr.a array radiator arrangement (equally spaced and nonequally
spaced) in the surface of the aperture, something which indicates the in-
variance of algorithm (2), and this means also, that of the corresponding
KOP w~tn respect to the location of the radiators. Tt likewise follows from
(6) that the properties of a ROAR junknown type of antenna array], which -
were studied in a eontinuous aperture approximation [1], are also extended ~
to a discrete aperture. Aowever, along with the cited continu3ty, certaiz
' specific features are also observed: -
2
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a) In the case of processing using algorittun (6), auxiliary algorithm (2)
and the KOP correspond~ng to it are not uniquely alike_ since one can take
as the filtering property of the d-functic~n in place of operator (2) in (6)
any other one:
i- . = J' f . . ~ - ~ .
. Joc, (R, K') cf~R ~ L .
in which �
y y y y
Jov~ (R,1~ = Joc~ (R,1C) ~PH R= R�
y ? y y ~ ~
Jov~ (R, ~ Joo~ (R, ~ nPH R ~ R� (7 )
tlie circumstance cited here points to the fact that algorithm (2) is re--
~undant with respect to operator (4), and th�s permits optimizing algorithm
L? and consequently, also the KOP within the limits of that freedom
in the selection of the amplitude-phase distribution which is permitted by
expressions (7);
b) The "Dirac comb" model in the form of an N-channel PVMS is an approximate
one, since each of its channels has a distributed structure and is not
described by the 8-function, but rather by finite width pupil function -
Jg(~, (for the sakP of convenience, this function is tied to the coordi-~ ~
nate system o� the antenna arr3y) and therefore, instead of (6) we h$ve:
N N .
LAP . = L ~ . . . .Ib ~Rr Rn~~ = ~j . . . Jav~ ~R? K~~ ~ LAP . ~8~
n=1 a=I
where Jopt (Rr, K') = L{J~ (R, R�)} ~ 1op~ (R,,, K') = L{S (R, R�};
As we see, the approximation of the "comb" wirh the pupil function results
in a distortion of the specified amplitude-phase distribution, and conse-
quently, in the r~quisite algorithm for coherent optical processing, and
- for this reason, it is necessary in a number of cases to accordingly compen-
sate for the pupil effect;
c) Inherent in the PVMS, as a system of N channels close together and con- .
trolled by an electrical field is the effect of mutual influence between the
channels.
Only the first of the effeccs mentioned here is studied in the following
using the example of a cylindrical ROAR.
A KOP was synthesized in i1, g~gure 1] which reproduces algorithm (2) for -
a continuous cylindriczl aperture of rad3:us Rp:
3
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~L . = F~ 1 {F . . J (Z)} 7' (2,~. S2z)}, (9)
where 10
T(~m~ ~z) = Fm {Jm (4) ~P iRo 1l f~2 - S2i cos q~)}, ~ 1
(10)
.7~~ ,1~ .7y ti~ .1y` ~ .
. i
r"u
r'V ^ ~ ~ _~r~ i 4~~,~,, ~ _
� � i~AA.~h~~ ~~i ~ ~ti~?
~ ~~l~~il?li~ 1~ ~ ~ .7,: ~I ~~~U~~I~~~~~ i S~
?~v
,J J??~~ b~I ~ v J'.I v v ~
Za d� Z%.
~
~ Ca) a
~c ~z /~j Q
y t,,.5~ ~S7y ///~~~T~ � S~� m
f
j~~
,~,R,R
H / xz ~K ~
' /
\ ~ .
/V ~ / ~ % ~ ` / /
~ ` /
p / / -N~~ :
; /i~// ~ . , .
/ ~ / ~
ii///~;:
~ i,
T, ' ,v ~ ; %
% _
~
jn i z 3 o,s;r
r,,,
- r /
~
v Q`
QZ _
T,~
T3n~ �
~ -
Q'' a~ T
~
Figure 1.
~ ~
F, F~ are operators for two-dimensional and one-dimensional Fourier trans-
forms respectiv~ly; K= 2~r/A, A is the radio wavelength; J(Z) and J~(�) are -
the partial amplitude-phase distributions; S2Z and St~ are the spatial fre- _
quencies. -
In accordance with (7), opeYator (9) permits a reduction. We sha'l1 demon-
strate the reduction using the example of simplification of transparency
(10), the structure of which is subsfiantially determined by the form o~ the
amplitude-phase response:
4
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Jv (~P, ~z) = Jm (41 exp iRo Y 1(` - S~Z cos ryj (11)
and consequently, is more susceptible to reduction. For a cylindrical
antenna array with radiators at the nodes of a rectangular grid (Figuse 1,
[1]), in place of (11), we form a new amplitude-phase response (Figure ia):
_ ~
Jm ~z) _ ~ Jm ~ndm, ~z) sinc ~~/~m - ii),; ~12~
n--ao
~ where sinc(n) is the readout function [2]; d~ = 2~r/N is the azimuthal
spacing of the grid (N is the number of elements in a ring). Since (12)
satis�ies (7), tfien instead of transparency (10), one can use a new trans-
parency (Figure 2b): .
T~ (52~, S>Z) - Fm {J~ (cp, SZZ)} _ ~ J~ (nd~, SZZ) F~ {sinc (~/dm - n)} _
n=--� �
~
= rect (52m/1~ N~ J(ndm~ S~z) exP (c~2~ndm) ~
n~--ao
~ .
= rect (S2a./]1n ~ 7' (S2m -F- nN, S2z)? .
~ 13 ~
where Poisson~s summii~g formula is employed j2]. As we see, in contrast to
transparency (13) is finite and entirely located within the rectangu].ar
window ;Figure lb):
N N
- - 2 ~S2mS 2 , -KSS2zGK, .
_ where for the sake of clarity, the reduction of only the real part of the
transparency transmittance is si;own, which is also shown in Figure 5 in [1].
We shall continue the reductior~ of the transparency, forming a new amplitude-
phase distribution of the following form (see Figure 1a) instead of (12);
~
J~o ~Z) _ ~ J~ -f- 2~cm, 52~,
_ (14 )
which likewise satisfies condition (7). For this reason, one can employ a
transparency~of the following form instead of (13):
T!~ ~ ~z~ = F,, {.i;~ ~Z)} = comb (~2~ Tt (S2ro, ~z)~
~ (15)
~
wh2re comb (S~m) S(52~ - n) is aii equally spaced "T~irac comb" .
n=-oo
5
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Thus, transparency (13) does not necessarily have to be reproduced entirely,
and it is sufficient to limit it to samples along the lines St~ = 0,
+1 +2, +O.SN (N + 1 lines in a11). Transparency (15) is weighted with
~he "Dirac comb" and is phys~.cally r.ot feasible. For this reason, we shall
approximate it with the ~o1low~n~ transparency (Figure 1b):
I
Tn~ ~Z~ _ ~ T 6 (S~ ~ n) 7' (rc, ~Z), ~ 16 ~ _
_ n=-~
where 1, S2~ = 0,
Te - p, ~52~ ~ ~0,5.
Transparency (16) is physically feasible, howeve~, in this case condition
(7) is violated. In fact, according to (10), the following amplitude-phase
distribution (Figure la) correspands to transparency (16):
~m ~Z) _ ~1 {T nt (52~, S~Z)} _ ~m ~~P~ ~Z). (17 )
_ where Jm (W) _ ~1 {T6 (SZ~)}.
If J~(~) = 1 when -~r/2 2 t .
6 ~ f=0 r
5 ~
c=2
4 ' -
J
/ ~
0 ~C ' ' ~ ~ . -
l,~ J 4 5 6 7 K 1~. 3 S S K'
In order to define K, it is necessary to obtain the distribution of the
lengths of overshooting. The figure gives examples of transition trees il-
lustrating the behavior of the sum above the threshold L if the accumulated
signals are uncorrelated. The transitions of the sum upward to s occur with
probability p or downward to 1 with probability q=1-p, where p is the pro-
bability of the appearance of the si~;nal s. In summing a number s~ 1, the
overshooting can begin with the state L+ where l= 0, 1, 2, ...,s-1. The
probability of the formation of overshooting wiCh a length of k steps from
li
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' the first entry of the sum into above-threshold states to the first entery '
into the below-threshold state is determ,ined by uumming the probabilities of
:.ill moves over the transition tree leading to t?~e termination of the over-
shooting at the k-th step. The summing of the ~robabiliti.es leads to the
formation of infinite series given in the table.
Searching for the rule of the formation of the tezms of the series obtained ~
on the basis of transition trees showed that the terms of the series caa be
~ compu~ed with the aid of the following normalized expression
(2)
k-l-l k-1-t sk~(11 �
W(k~ ~ k~ Crs+t p s+t 4 s+i ,
where g=(~+-~)~ (1+1)+(~rs)~ (~+1)+~2(I+s),...,(lil)~m(~+S)~...~''''.
I I � k
a I ~ I ~ I 3 I 4 I b I g I ~ I 9 I 9 I 10...
I , : . ~ ~ p 14 ~ � 0...
l 0 I q I ~ I/'9 I ~ I2p9 I~ I5P9 I I PQ I
I
I
~ Q I p I p I Fqa I p I p I 3p+qs I 0 I 0 I 12P 9
2
] I 0 I q, I 0 I 0 I 2P4. I p I p I 5 p'q' I 0 I 0.
~
By using the induction method it is possible to show that expression (2) is
applicab~.e for s~ 0. When s= 0, the distribution (2) changes to the Pascal
law, The distribution W(k) is obtained from (2) by averaging~over a random -
value As a r~sult of weight addition of nonoverlapping lattices W(k,~
the W(k) distribution also is a lattice in which nonzero terms are obtained
: when k=(l+s)(1-kn), where m~, 1, The average length of over-
s~hotting is aqual to
k=~ ~_i t=s_~ ~3 )
ti = L '~~(k) ~ ~ k ~ ~t) ~ ~~l)~
k-1tl 1=0 !=0
where
k= ee .
� k i~) _ ~ ~i~ (k, 1). ~4~
c=t�+t ~
Direct computation b~ (4) shows that K(~ )=(1+ ~)/[1-p(s+l)]=1+ c~)/(q-ps),
s 7 0. The value of R is determined by the numerical method.
In order to determine the probabilities W(n ~ L) and W(~), it is necessary to
find the distribution W{n) of the sum in the TsRN. The W(n) distribution can
be calculated through solving the matrix equation [3]
16
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_ ~
lT'(n)(B-I)~-0, ~5~
, wl~ere B is the matrix of the probabilities of transitions of the sum; I is
the iinit matrix.
For the ergodic Markov ciiain, solution (5), which can be converted to an al-
gebraic system of difference equations with constant coefficients has the fol-
lowing form when s=1:
l~ (n) ~ - q / \ 9 ~R~ ,
and for s~1, the solution can be expressed approximately through the expon-
ential rule. For the exponential rule of W(n), average lengths and the re-
currence frequency of noise overshooting are equal to:
K s]!~9- ps)~ Fs - Fn qW (n m l.)� ~6~ _
Expressions (6) are exact at s=1 and are approximate at s 7 1. The simu-
lation of TsRN on a digital computer by the Monte-Carlo method showed that
at s L 4 the accuracy of the determination of the value F g by (6) is not
worse than one percent. It can be shown that the expression for F~ at s=1
remains exact even if the capacity of the reversible counter is limited,
which results in the limitation of the amplitude of overshooting, and that
at s~ 1 its accuracy, as was shown by simulation, is retained regardless of
th,~ capacity of the counter.
Bibliography
1. Minguzzi, B., and Picardi, G. ALTA FREQUENZA, Vol 39, No 11, 1970.
2. Likharev, V. A. "Tsifrovyye metody i ustroystva v radiolokatsii" [Digi-
tal Methods and Devices in Radar], Sovetskoye radio, Moscow, 1973.
3. Kemeni, Dzh., and Snell, Dzh. "Konechnyye tsepi Markova" [Finite Markov
' Chains], Nauka, Moscow, 1970.
COPYRIGHT: Radiotekhnika, 1979
10,233
CSO; 1860 -
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UDC 621.391.2
EPXERIMENT~L STUDY OF SPACE-ZWO-CHANNEL DETECTION RECF.IVER--INTERFERENCE
COMPENSATUR
Kiev IZVESTIYA VYSSHIKH UCHEBNYKY ZADEDENIY: RADIOELEKTRONIKA in Russian
No 7, 79 signed to press 10 May 78; after revision 10 Oct 79 pp 98-11)0 -
~Article by B. A. Lszutkin, A. B. Lazutkin, V. V. Mansurov, B. B. Pospelov~
~Text ~ In ~1~ wa3 obtained the structural arrangement of an opt,imal
receiver with respect to the criteriori of the maximum ratio of probability,
and a two-channel version of such a receiver was studied analytically,
The possibility was shown as, a result of such a study, of the possibility
of full compensation of stationary interferences, created by an external
source, located at an srbitrary point of a remote zone. It was assumed _
in the analytical study that averaging multipliers (correlators), multi-
pliers, band filters, summing and subtracting devices are ideal and,
further more, phenomena of limiting interferences and useful signals in
various receiver components were not taken into account. The characteristics
of actual devices differ from the ideal ones, therefore, to check the degree
of compensation of interferences in a space-two-channel receiver ~1~ ,
an experimental device was assembled, with a structural arrangement as
shown in Fig. 1, w}iere G4-lA1,G4-1A2 are standard signal oscillators used
to initiate the hi~;h frequency voltages of interferences and the useful
signal at the outputs of a two-channel antenna with phase channel centers
spaced by value d; G2-37 noise oscillator to produce amplitude modulation
of the output voltage of oscillator G4-lA, by noise; M1, M2 pulse
modulators of interference and useful signal voltages respectively; ~
regulated phase rotator used to provide a phase difference
~ dsinA
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between high freqt~ency interference oscillationa in the first and second -
= channele af the receiver angular dieplacement of the intarference
source with resr.ect to the reference direction of the two-channe~ antenna);
C1-54 monitoring oscillograph; EP emitter repeater to provide
matching of modulator M1, M2 outputs to inputs of the two-channel receiver;
PRM space-tw~o-channel receiver (without antennas) in which, instead of
single-dimensional filters matched to the spectrum of the useful signal,
single-dimer~sional filters matcheu to the width of useful signal are used.
(1) ' (2j - -
~4-1A M~ 3!1
(4) .
~ 3~ 2:3 P 9yK C/'.yi
~~-iA M 3n
Fig. 1
1. G4-lA 3. G2-37
2. EP 4. PRM-VLTK
The PIrM operation ie shown in Fig. 2~l~ .
,
f ~
~ .wt P:;
~'ig. 2 -
19
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The PRM was assembled with ordinary parts used in household equipment.
For example, phase detectors are used as averaging multipliers and broad-
casting receiver fil.ters are used as band filters, retuned for the working
frequency of the installation fo = 500kHz. The following signals were
sent to the inputs of the two-channel PRM (Fig. 1) during the experiment:
1) useful signals in the form of a sequence of simple radio pulses with a
frequency of high frequency oscillations equal to f=500kHz, pulse
length considerably smaller and the period of~following TN
considerably reater than the time constants T2 of the averaging
amplifiers ~1 ~ ('C~ ., Z=~ 1, X= x~,t) = Wi;("~ t):
. r
~7~ aN'~=Lw~~(~,t)+~v{,-~,)Wa~~~,t)+~ ~n,~~'~~(~~t)~
Here the coefficient ~ ~ is determined from the condition of normalization
m ?
~ ~ ~ J vt~~r~ t) d~,.
t~s f-s
By adding up the left and right sides of the equation (7) with all possible
values of i, j it is not difficult to obtain the equation for unconditional
a posteriori density of distribution W(~ ,t), of the continuous parameters:
aN'~~'~t~ - Lw(a,, t)+[zr-~o]i~(~,
at~
~ where
m 1 (Atf)s=
Zx = Tu ~~~pri exP [ f Fcrdt] -1}, _
ks�
pi, = p(Z i, X= x~) is the a posteriori probability of states Z=~ i,
J
X= x, and the notation F., = F(~ Z=~ X= x, t) is introduced.
J i~ ' i' J ~
Strictly speaking, the probabilities pi~ have a conditional nature, but for
problems of structural synthesis it is possible to restrict the case of
independence of pi~ from parameters ~(5).
Considering the a posteriori density W(~ ,t) which is valid for the condi-
tion of applicability of Gaussian approximation, it is possible to convert =
from equation (8) to equations for the estimated values of the com-
ponents of vector ~ and of the c~nulants h~~~ :
26
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~ (10) d1~d =aa(~,',t)-f-~hoa 8CV>z:'
dt 8~,a' '
a_s
(1~) dhaA~ aaaa~.� t> h"~+ aad
r ha~~+ _
dt r
Naa n 8~0,
r_t -
where Pr~l~ is the probability of accurate relay and Pr~~~ is the probability
of a certain error in relaying a train with r repeaters.* Detection of
the synchronizing pulse and each of the information pulses is accomplished
independently. In this case OAOKP coincides in essence with the algorithm
for known from the theory of multiple-alternative signal detection in noise
(1), namely, the synchronizing pulses hold position i corresponding to
m
(2) n~aiccI~p~~n~t)/po~n.r) ~i=1,2,...,.N),
,_t
while position js of the information pulse s corresponds to
. M$XCpi ~nm}a,~~ ~~110 (nm+..i~ ~ r 5=1~ `1.! . . . r l 1 ~
j \j.=1,2,...,Nl
,
*In communication systems of practical interest~ Prl~ Pr~~ which is why
(1) is done.
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where p~(n) and pl(n) are the probabilities of recording n photoelectrons
at time j, in the absence and presence of a signal respectively. ThusA
in Implementing condition (1) OAOKP does not depend on the number, struct~irc
and nature of the repeater operation. With m= 1~ (3) coincides with (2);
for this reason only algorithm (2) is considered below.
z
' t
0 r T T 2T..... YmTz'f~ (m+11T ~p..... ~t (m+!)T
N~nyabcbi cu~spoHU3aur~r~ Ny~opMam~e,vaie unny~ecd~
Figure 1. Transmitted Pulse Train
Synchronization Pulse Information Pulses
OAOKP (2) does not determine unambiguously the structure of the receivers
in the repeaters, but since this structure has a significant effect on
the noise-immunity of the entire line's reception, it is necessary that
the receivers in the repeaters provide maximum noise-immunity for the
communication line.
The probability of accurate reception at the end of the line equals
n
~4) Fapa~=F+~ P~''
where F1 is the conditional probability of accurate reception of the periodic
train assuming its correct relay and F contributes mutually compensated
errors to F~p~t~ . Analysis of formula (4) shows that the requirement for -
maximum noise-immunity of the reception of the entire line makes it im-
possible to implement receivers in the repeaters b~~~d on algorithms
which are optimum in the sense of having maximum P . In this case it
may be calculated with an adequate degree of accuracy that
R
' . ~5) Fnree=F'~~ P~'i�
,r~
Specifically, if the transmission to the repeaters is doen in the optical
range, reception at the repeater must be performed by (2) with P~1~ calculated
the same as F1. r
The probability F1 is equal to the probability of the inequality
(g) ~,("nm~)>~,~n~u), _
49
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where
nm~ _ {n~{, nz{, . . . ~ nm~} ~ ~ ~~mr) _
~
~ p~ ~n.r) ~Po ~n,t),
,_t
w
and the values ~i~ and n~ are only caused by noise and signal plus noise _
respectively.
~
Inserting the set D~nmi~~ ) of values nmk which satisfy (6), we write F1
in the form �
, r N-i
(7) F~ _ ~ (itCms ) I y ~ Po ~nmk~ 1 ~
i
rimi�~ ~ nmkED~T~Mni ~ .
where
. m TM _
' Po (nmm) _ ~Po ~n,~) ; F's ~nm~~) P~ ~n.~) �
. ,s~ ,-s
~ ~ .
Addition by n in (7) means addition by all m coordinates of n u' in the
range of the set D. Addition by the first coordinate of n~u'results in
from 1 to~ an~l by the other coordinates of r~~"Sresults in from 0 to
The optimum spatial distribution of the repeaters may be ascertained by
solving the system
- (g) aF~D~ =0,' r=1, 2, . . . , R,
. a~r
where x is the distance~between adjacent repeaters. With a fixed length
of the entire line X=~ xr of system (8) virtue of (5) wili take the
f orm r : i
aP;'' aF,
(9) ax* F,-Pr =0; r=1, 2, R.
a~R+~
Solving (9) we obtain the values xl,xZ,...,xr which yield the maximum F;rpc~e
which we designate F~~T,~. If FoPrR does not reach maximum in terms of R,
the number of repeaters for reliable operation of the communication line
must be looked for from the condition L+
. ' FOas,RiL' pf
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where F~ is ttie assigned value of noise-immunity of reception. If all ttie
transmitters of the system radiate in one and the same range and liave equ~il
power and the properties of all the intermediate channels are identical, _
then the uniform distribution of t e re eaters (x} = x2 =...xr = X/R+~)
is optimum, being in this case pi1~= p2~)= =PR1~ = F1'
To obtain quantitative results it is necessary to know.the principles c~f
distribution of the noise photoelectrons pp(n) and the convolution of
~ the 3istribution of signal and noise photoelectrons pl(n). A~ is known, _
the problem of calcuJ.ating the different kinds of noise in optical commimi- -
cations has not been solved (2); the form pp(n) and pl(n) is found only
_ in a number of limited cases. Inasmuch as a rather general problem is
considered here (spe.cifically, open and enclosed lines). we are limited to
detecting the greates*_ ancl least of the possible values of reception noise-
immunity for arbitrary co;~r.nunication lines and the noise properties of a
photoreceiver (as regards the photoreceiver~ it is assumed only that it
operates in a mode such that the single-electron pulses are discernible)e
For this purpose the calculations are performed at two extremes in the
sen~e of deviation in the distribution of the cases: for Poisson
distributio~i (PR)
po~~)= ~"(n!)� BxP~_'"(IIi)? P~~n)= .
t10~ ~~^(~-~"1'c)~)�
= exp (-~~-~o) T,
n!
which has the least deviation and for the distribution
Po ~n) _ ~"(~ti) " (,~mti~ n
~1+'~mz) n+t ; P~ ~n) _ ~1+T~~~ X
- ~(11)
-'Y~i
Xexp ~ Ln ( 1 ,
1+'Y~i ~'Ym(i+^(~i) /
which has the greatest deviation and repres~nts the convolution of a single-
mode Gaussian noise with a Poisson signal (2) and Y~ in (10) and
- (11) are the average intensities of the noise and signal photoelectrons;
I.n(x) is the Laguerre polynomial). As is shwon in (2), other variations of
the distribution including multimode sound radiation, shot noise and so on
have intermediate deviation in comparison with (10) and (11).
In (10), algorithm (2) stands for detection by maxim~im cumulative number of ~
photoelectrons m
- '~12) ~ n,~=hiaxc,
,o, '
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and, due to the additivity of the Poisson distribution, (7) is considerably '
simpJ_ified:
m ~ ~'Y~~',~~) mz) n
~(13) F,=~ n~ exp(-~o,-~~)mTX
~ n-! /�~~111S~k N i
X `_---e:cp(-~Wmi)~ .
k!
k~0
_ In (11) algorithm (2) has the form
m
(14) ~lin,~ 1 - ~e ~ =11i8RC
~ NW (1+^(mi) ~
._t -
and with m= 1 i.n virtue of the monotony of I.n(x) with respect to n, (14)
also signifies reception with the maximum number of photoelectrons. In
(11) inequality (6) has the form
m m
- ~ L (x) ~ ~ Lk ~
s=1 n8 e=1 ~
where
x=-~~i (1+1(~i) �
Inserting the function x= Un(y), in inverse ratio to y= Ln(x), and calcu-
" lati.ng that L~(x) = 1, we write (7) in the form
- 1 - y~mt
(1.5) Fl = (1 -f- Vmi)t"N e~p ( 4 ) X .
m
~ rID~ 1 ~ rta i'.(
_ X~~... L` IB-1 1 j Ln' (x~ X
1 '~~t f e_1
711=197==~ 71~-~ ~
m n?
, Uk~(n Ln~-1 U~(II LnlLk~~-1
s=1 D 8=1 �
X k~ k~ -
U
m m-1 \ h-1
Ukm Lne / n Lk8 I-1 , m
_ \a=1 8=~ / ~ ~ ~k
Ym 8_1 ~
~ . ~ ( 1 ,~m,~ - '
km 0
The optimum reception algorithm (2), practically unrealizeable due to
52 .
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equipment difficulty, may serve as the standard of quality for quasi-
optimum algorithms.
Reception with the maximum number of excesses of a certain threshold n,
when the position of the pulses of the train is determined from the con-
- dition
m
gr= ~ x ~n,~) n.r=a~axc,
- ._s '
where
x(n)=~0, nn.
is a conveniently implemented version for simplification of the optimum
algorithm (12).
Essentially, the numbers gi differ from zero only for a small number of -
positions il,i2,.~.~ik and the number of these positions k is an arbitrary
- magnitude the numerical distribution characteristics of which, according
to (3), satisfy the inequality
~
- 6~a.
1
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It is evident that there exists a definite dependence between An and Q(,n,
therefore, having expressed Ql = f(A), it is sufficient to perform averag- -
ing only over 9n. Since the time delays n of reflected waves are inde-
pendent, and their root-mean-square deviation on real routes are greater than -
the period of oscillation of the metric and decimetric ranges [5] then it is
possible to consider that the random phase angles (Pn = C,Jo 2n are statis-
tically independent from one another and of An. Then the right part of (4)
can be represehted in the form of two separate averages over An and .
N
~EorptEo~pi> � 0,25Ei ~1=i ~u is
~o ~ /0~
~ 0,15 Q5Q 475 1 er,)eT 0 1 a? 3 rry
Fig. 1
1. P~~Pw~ 2. db
It follows, therefore, that there exists an optimal value of the threshold
level of the analyzer ~,t which the noise resistance of the receiver will be
maximal for two kinds of noises.
It was assumed previously that the length of equal frequency intervals dZ
- in the equal signal zone is constant and is equal to the average interval
between the blips of the naxrow band pro~ess. This limitation may be remove3
if the distribution density of such intervals W( Qt ) is Imown. Then~ instead
of (3), it is necessaxy to write
m ~
Po (T) = ~ ~ i - I~ez~ w ~e~~ dez
~s~
and ma.ke similar calcula.tions. Regrettably, no expression fo W(6T ) conven-
ient for calculation has been published even for a Ra,yleigh model of fading.
The relationships obtained in the paper may be used also for evaluating the
actual noise resistance of an automatic selection installation when organizing
the usual space-diversity reception. In this case, the reception frequency
is constant P(i2,)=P(s~2)=~ and multiplier 0.5 in fron.t of ~oeffi-
cient Y disappears in expr~ssion E6). Then, for a full inphase stat~ of
signals in ~ranches ~.Y�=1 independently of threshold level
pc
-?oo,
pmK
which was to be expected.
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- BIBLZOGRAPHY
1. Sakata Tatsuporr. Communications system far running trains. Jap. Rail- ~
way IIzg.~ 1970~ 20~ No 10, p 25.
2. Bilenko, A. P.; Morgunov~ L. N.; Kuz'min, V. I. "Multichannel System of
the 'Altay` Mobile System of the Ultra-Shortwave Ra,dio Communications."
"r~,EKTROlSVYAZ' . i 972 ~ No i, p i.
_ 3. Zyuko, A. G. "Interference Resistance and Efficiency of Communications
Systems." Moscow, "Svyaz'," 1972�
4. Baxsukov~ Yu. K. "Correlation F~uiction and Spectrum of Sinusoidal Oscil-
- lations for 5udden Random Changes of Frequency and Phase. " RADI(nIICH-
xIKA I ELIICPRCNITCA, i965~ 10, No 4, p 595�
5. Tikhonov~ V. I. "Blips in Random Processes." Moscow~ "Naukat" 1970.
6. Golovin, E. S. " Qi Antenna Effect on Sta,tistical Chaxacteristics of the
Signal ~hvelope in Communications Systems with Mobile Objects."
RADI(nIICHNIKA, 1977~ 32, xo z, P 3~� ~ .
. COPYRIGIir s "TZVESTIYA WZOV SSR - RADIOII,EKTRINIKA" ~ 1979
[269-2291] -
2291
CSO: 1860
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I
- ELECTROCOMMUNICATIONS EQUIPMENT AND ITS PRODUCTION
Moscow APPARATURA ELEKTROSVYAZI I EE PROIZVODSTVO in Russian 1979 signed
, to press 25 Oct 78 pp 2, 326-328
[Annotation and table of contents from book by Anatoliy Filippovich
Gryzlov; Aristid L'vovich Meytin'sh; and Yuris Petrovich Pone, Iz~atel'stvo
"Vysshaya shkola," 17,000 copies, 328 pagesJ
[Text] The technology of assembly and wiring operations during production
of electrocommunications equipment is examined in the book. Particular
attention is devoted to the switching and electroradio elements which form
the basis for switching and channel-forming equipment. Infoimarioz on
telephony and telegraphy, as well as the principles of multichannel communi-
cations are presented, and power-supply devices are described. Individual
chapters are devoted to interchangeability. tolerances and play, technical
ar.d electrotechnical measurements, quality control and labor safety
procedures.
Table of Contents Page
Introduction 3
Chapter l. Tolerence and TEchnical Measurements 5
1. Interchangeability of parts 5
2. Tolerances and -lay g
3. Dimension chains 12
4. Technical measurements 15
Chapter 2. Technology of assembly operations 20
5. The industrial process 20
. 6. Assembly of separable ar.d non-separable connections 2S
7. Assembly devices 2g
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Chapter 3. Technology of wiring operations 31
8. Wire installation 31
9. Printed circuit installation 34
10. Construction of printed circuit boards 38
11. Technological foundations of printed circuit boards 41
12. Technology for producing printed circuit boards 46
13. Electrical c:~iring connections 48
14. Soldering electrical wiring connections 54
Chapter 4. Electroradio elements making up a set 62
15. Resistors 62
16. Condensors 64
17. Electron, ion and semiconductor devices 66
18. Integrated microcircuits 69
- 19. Items with coils 73
20. Auxiliary elements 76
Chapter 5. Switching elements 80 ~
21. General information on contact systems 80
22. Contact springs 83
23. Contact groups 86
24. Magnetically controlled contacts 89
25. Mechanically controlled switching devices 92
26. Selectors 96
27. Neutral electromagnet relays 98
28. Folarized relays 104
Zy. Snap contact relays 107
30. Multiple coordinate connectors 112
Chapter 6. Techiiology of manufacturing switching elements 115
31. Manufacturing contact groups 115
32. Assembly and adjustment of the RPN relay 119
33. Assembly and adjustment of the RES14 relay 123
34. Assembly of relays with magnetically controlled contacts 126
35. Technology of switching unit assembly 12~
Chapter 7. Fundamentals of telephony. Telephone sets 134
36. The concept of sound and audio perception 134
37. Principle of telephone transmission 136
38. Parts and assembly units of telephone sets and ,
electrocommunications equipment 140
39. General purpose telephone sets 148
40. Pay telephones and other telephone co~nunication -
equipment 155
98
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Chapter 8. Telephone switchboards 159
- 41. General information 159
42. Schematic diagram of the URTS-100/600 switchboard 161
43. Schematic diagram of the M-60 switchboard 169
44. Long distance MRU [Interrayon center] telephone
e~Efice switchboards 175
Chapter 9. Automatic central telephone offices 180
45. Principles of automatic switching 180
46. Simplifies schematic diagram of the ten-step system 187
47. Crossbar system telephone offices 194
48. Quasi-electron automatic telephone offices 199
Chapter 10. Electrical metering techniques. Control-and-
measuring equipment 203
49. Metering in communication practice 203
_ 50. General information on electric meters 205
51. Metering current and voltage 208
52. Measuring resistances and capacitances 211
53. Oscillographing electrical oscillatory processes 215
54. Instrument generators and frequency measurement 217
Chapter 11. Multichannel communications equipment 211
55. General information 221
56. Audio frequency channels 223
57. Telephone amplifiers 227
58. High-frequency channels 228
59. Generators 234
60. Modulators 238
61. Systems for multiplexing communications circuits 242
62. Multichannel communication systems with pulse-code
modulation and time division of channels (IKM-VD/PCM-TD/) 249
Chapter 12. Telegraph communications 254
63. Principles of telegraph communications 254
64. The STA-M67 start-stop telegraph set 258
65. The AT-PS-PD automatic telegraph station 277
Chapter 13. Electrical power supply for communications equipment 283
66. Electrical power sources 283
67. Rectifier devices and installations 287 -
68. Power supply installations for automatic telephone
offices 291
99
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69. Power supply ~-~stallations for lc:ig distance communica-
tions enterpr~ses 293
70. Power supply installations for telegraph stations 296
Chaptc~r 14. '1'cchnlcal in5pcction 29~
71. Quality of electrocommunication equipment items 298
72. Organization of quality control 301 -
73. Testing i.tems 305
Chapter 15. Mechanizing and automating production 311
74. Basic concepts and definitions 311
- 75. Using machinery and automatic equipment 313
Chapter 16. Labor safety procedures in the production of
electrocommunications equipment 316
_ 76. Safety techniques 316
77. Electrical safety 319
78. Fire prevention techniques 322
Literature 325
COPYRIGHT: Izdatel'stvo "Vysshaya shkola," 1919
9194
CSO: 1860
100
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Converters, Inverters, TranGducers
UDC 535
ELECTRONIC-OPTICAL CONVERTERS AND THEIR USE IN SCIENTIFIC RESEARCH
Moscow ELEKTRONNO-OPTICHESKIYE PREOBRAZOVATEL I IKH PRINIENENIYE V -
NAUCHNYKH ISSLEDOVANIYAKH in Russian, 1979 pp 432
[Annotation and table of contents from book by Mikhail Mikhaylovich Butslov;
Boris Mikhaylovich Stepanov; and Sergey Dmitriyevich Fanchenko, Izdatel'stvo
"Nauka," 432 pagesJ
[Text] This book considers a broad calss of devices for spectral conversion,
amplification and analysis of pictures. The first section presents the
physical principles of forming and tramsmitting pictures. It also con-
siders certain general properties of pictures with brightness near the
threshold and of short duration. The second section of the book describes
electronic-optical converters and brightness amplifiers of various systems,
while the last two chapters of the fourth section describe experimental
apparatus and apparatus manufactured by industry for electronic-optical
high-speed photography.
The third and fourth sections present a.detailed review of the use of elec~-
tronic-optical devices and apparatus in physical reseaxch (including laser
and thermonucleax), in astronomy, biology and med.icine.
Table of Contents Page ~
F`rom the ed.itor (
Foreword g
Introduction li
~ V.i. Brief outline of the development of the equipment
and the use of EOP (Electronic-optical converters) 11
V.2. Principles of picture transmission 14
y~3� Obtaining electronic pictures in an electrical field 18
V.4. Problems of picture metrology 22 -
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Table of Contents
Section 1. Ce~~tain General Properties of Pictures 27
i.l Fermi-H~; lton principZe 2~
Chapter 1. Aberr~tion of electronic pictures 30
1.2 Geometric '~,berration 30
1.3 Chromatic ~~berration 38
Chapter 2. Time res~lution of pictures 43 .
1.4. Physical time`,resolution of light optics 43
1..5. Physical time i~solution of electronic-optical converters 47
1.6. Time resolution~~~f equipment 55
1.7. Certain conclusio~s on time resolu~ion of optical _
and electronic-opiFical systems 60
Chapter 3. Informational properties of pictures 62 -
i. 8. Qua.lity of pictures 62
1.9. Contrast-frequenc~ characteristic 63
1.10. Methods for determining the contrast-frequency
chaxacteristics 67
1.11 Fluctuation properties of pictures.
Generalized quanium output 70
1.12. Transmission of brightness~g~adations of. pictures
.Information concept 76
1.13. Information capacity anci information quantum
output of an ideal ima,ging device 79
Section 2. Electronic-Optical Converters and Picture
Brightness Amplifiers 85
Chapter 4. Focusing systems for electronic-optical converters 85
~ 2.1. Electrostatic immersion lenses for EOP 85
- 2.2. Magnetic focusing EOP systems 92
Chapter 5. Semitranspaxent photdcathodes with an external
photoeffect 96
2.3. Silver-oxygen-cesium photocathode . 97
2.4. Antimony-cesium photocathod.e 104
2.5. Multialkaline photocathodes 109
2.6. Structure of semitranspaxent photocathod.es 113
2.7. Photocathode transfer i16
2.8. ~rther improvement of photocathodes ii9
2�9� Distribution of initial velocities of photoelectrons 125
Chapter 6. Luminescent screens for electronic-optical
converters 130
2.10. Investiga,tion of light output and spatial resolution
of sulfide and sulfide selenide screens 130
102
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- Table of Contents
2.11. Methods for manufacturing luminescent screens
with high spatial resolution 133
2.12. Light chaxacteristics of luminescent screens i36
Chapter 7. Ba,sic types of electrontc-optical converters 141
2.13. EOP with direct transfer ;i pictures 141
2.14. Time-analyzing EQP 149
2.15. Electronic-optical dissectors 166
2.16. Amplifying EOP with magnetic focusing of picture 167
2.17. Phot ocontact EOP 170
2.18. Electronic-graphic picture converter 172
Chapter 8. Picture brightness amplifiers 181
2.19. Multichamber EQF with ma,gnetic picture focusing 181
2.20. Cascade secondary-electron amplifiers of picl�.ure
brightness i88
2.21. Nfultichamber EQP with electrostat'-.~ picture focusing 189
- 2.22. TV transmittin~ tubes with amplifying taxget 192
2.23. Microchannel amplifiers of picture brightness 195
2.24~. Amplifiers of X-ray pictures 199
Chapter 9. Noises, sensitivity and informational
chaxacteristics of picture brightness amplifiers 205
2.25. int ernal electronic noises of picture brightness
amplifiers 205
2.26. Dispersion amplification coefficient and generalized
quantum output of multichamber EOP 214
~ 2.27. Amplification of picture brightness 217
2.28. Contrast-frequency chaxacteristics of electronic-
optical devices and their ba.sic camponents 219
2.29. Information capacities of EOP and emulsions 224 ~
Section 3. Scientific Application of Amplifiers of
Picture Brightness 227
,
Chapter 10. The use of amplifiers of picture brightness
in nucleax physics 227
3.1. Luminescent cha,mber 227
3.2. Spaxk counters with subnanosecond resolution 255
3�3� Photc~aphing Cherenkov's radiation of relativistic
chaxged particles 257
3.4. Rec ording neutron pictures, as well as tracks in
streamer chambers 2b9
Chapter 11. The use of brightness amplifiers in optical
- spectroscopy and plasma physics 272
3~5. Met hod of electronic-optical spectrochronography 272
3.6. Spectroscopic investigation of p~asma. using EC~ 277
3~7� Qptical spectroscopy of accelerator beams 282
103
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Table of Contents
Chapter 12. The use of amplifiers of picture brightness
in astronomy 284
3.8. Threshold sensitivity of astronomic observations 284
3~9. Astronomic installations with brightness amplifiers 291
3�10. Astronomic results obtained by mea.ns of amplifiers
of picture brightness 300
Chapter 13. The use of picture brightness amplifiers
in biology~ medicine and microscopy 314
3.ii. Micros~opy with picture brightness amplifiers 314
3�12. X=ray and autoradiography with the amplification
' of the picture brightness 319
3.13. Results of using brightness amplifiers in medical X-rays 325
3.14. Microbiological results 3z7
- 3.i5. Measurement of quantum output of the eye 329
3�16. Achievements in medical autoradiography and in
structural X-ray analysis 330
Section 4. The use of E~ in The Investigation of Fast
Processes 33z
Chapter 14~. F~cperimental electronic-aptical installations
for investigating process in the nanosecond
and subnanosecond ranges 332
4.1. Single-frame electronic-optical cha.mbers 334
4.2. Multiframe time loops 337
4.3. Installations for electronic-optical chronography
with continuous picture scanning 341
4.4. Installations for electronic-optical chronography
with pulse linear picture :~canning 3~9
4..5. The use of picture brightn~ass amplification in
oscillography 361
Ghapter 15. Industrial electronic-optical apparatus for
high-speed photography 3~
4.6. US-01 installations and series LV time loops 3~
4.7. Series LVE time loops 3~
4.8. Chambers for series FER electronic-optical chronography 370
4.9. High-speed electronic-optical chambers "KA,DR-4-ZIS~" _
"KANAL" and LV-05 376
4~.i0. Industrial high-speed electronic-optica? chambers in
the Unite3 States, Great Britain and France 377
- Chapter 16. Investigation of the nanosecond and picosecond
processes by the electronic-optical method . 382
� 4~.11. Determination of the real time EOP resolution 38Z
~.12. Investigation of spaxk dischaxges 39~
4.13. Laser investigations 396
Bibliography 447
COPYRIGHT~ GLAVNAYA RIDAKTSIYA FIZIKO-MATEMATICH~',SKOY LITERATURY IZDATEL'STVA,
"NAUKA"~ 1978
~27'2291~ FOR OFFIC~~ USE ONLY
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621.317.4
ELECTRICAL CONDUCTIVITY SENSORS -
Moscow DATCHIKI ELEKTROPROVODNOSTI in Russian 197~ signed to press 19 Dec 78 _
pp 2, 156-167
[Annotation and table of contents from menograph by Mikhail Matveyevich
Zakharov, Izdatel'stvo "Nauka", 1800 copies, 156 pp]
[Text] This menograph discusses sensors based on inductive windings and
ring magnets and also sensors in the form of systems of electrodes of
various conEigurations. A theory of sensors is presented, taking account
of the nature of interaction of a quasistationary electromagnetic field
with a semiconductor, and problems of analysis of sensor characteristics -
are solved.
- The book is intended for scientific, engine.ering and technical workers.
Contents Page
Introduction 3
Chapter 1. The Electromagnetic Field of a Ring-Shaped Circuit
Containing Magnetic Flux and a Turn Containing
Magnetic Flux 7
1.1 The Helmholtz equation for the vector potentials of an
- electromagne~ic field 8
1.2 The vector potential of the field of a circular magnetic
- circuit located above a layered semiconductor medium........... 11
1.3 The vector potential of the field oE a circular magnetic _
field placed above semiconductors in a two-layered medium,
a sheet and a half-space 16
1.4 The vector potential of the field of. a current-carrying
loop placed above a layered semiconductor medium............ 18
1.5 The vector potential of th~ field of a current-carrying
loop placed above a semiconducting two-layer medium,
a sh~.:et and a half-space 21
1.6 Determining the electrical and magnetic field intensity.......... 22
1.7 The interrelationship of electromagnetic fields. The
vector potential in free and semiconductor space 24
105
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~ Chapter 2. Surface-Applied Inductive SenSors 29
2.1 ~utput characteristics of sensors ua~ed inductive coils of
negligibly small cross section 29
2.2 Output characteristics of sensors based on solenoids 35
2.3 Output characteristics of sensor.s based on thin-walled
~ cylindrical inductive coils 38
2.4 Output characteristics of seii~ors based on inductive _
coils of rectangular cross section 43
2.5 Replacement schemes and calculation of output character-
istics of sensors based on inductive coils of rectangular
cross section 46
2.6 Main laws of the operation of surface-applied inductive
sensors 49
Chapter 3. Surface-Applied Capacitive Sensors S8
3.1 Output characteristics of sensors based on circular
magnetic circuits of negligibly small cross section...~........ 59
- 3.2 Ot~tput characteristics of sensors based on circular
thin-walled magnetic circuits 62
3.3 Replacement schemes and calculation of input characteristics
of sensors based on magti(�tic circuits of rectangular
cross section 66
3.4 Main laws of operation of surface-applied capacitive sensors..... 70
Chapter 4. Through and Imbedded Sensors 74
4.1 Vector potential of the field of a circular ma~tic
circuit encircling a semiconducting cylinder 75
4.2 Vector potential of the field of a circular magentic
circuit surrounded by a semiconducting medium 80
4.3 Output characteristics of capacitive sensors 84
4.4 Vector potential of the field of a current-carrying loop
encir~ling a semiconducting cylinder 87
4.5 Vector potential of the field of a current-carrying loop
surrounded by a semiconducting medium 89
4.6 Output characteristics of inductive sensors. Some results
of numerical calculations 91
Chapter 5. Electrode Sensors 96
S.1 Determining the potential functions of electrodes of
various configurations 98
5.2 Calculating the parameters of sens~rs with cylindrical
electrodes 106
5.3 Some results of numerical calculations and simulation of
sensors 113
Conclusion 120
106
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- ~ Appendiceg -
I. Analytic anc; numerical methods of calculating improper integralA.. 121
� II. Calculating the field of a circular magNetic circuit placed abov~
a semiconducti.:~g half-space 127
III. Calculating the field of a circular magnetic circuit placed
above a semicon~ucting sheet........... 130
IV. Calculating the f~:.eld of a current-carrying loop placed above
a semiconducting half-space 131
' V. Calcul.ating the field of a current-carrying loop placed above
a two-layer medium 133
~ Tables 135
' Bibliography 152
COPYRIGHT: Izd3tel'stvo "Nauka", 1979. -
8480
CSO: 1860
107
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- Electrical Engineering Equipment and Machinery
UDC 621.371.5.[Illegible] 11 . _
MATHEMATICAL SIMULATION OF EXTERNAL ELECTROMAGNETIC FIELDS OF SOURCES
Minsk IZVESTIYAVYSSHIKH UCHEBNYKH ZAVEDENIY: ENERGETIKA in Russian No 6,
1979 pp 34-38
[Article by Docent S. M. Apollonskiy, Candidate in Technical Sciences]
[Text] ~1t the present time various methods of passive shielding are used for
lowering external electromagnetic fields of electric equipment. The deter-
mination of electromagnetic fields and calculation of shields present cer-
tain difficulties.
However, the solution of problems of shielding can be simplified substantially
if the real sources of electromagnetic fields are replaced by their equivalent ~
models which are a set of dipoles. As was shown in [1], it is possible to
create them if there are available data about the nature of the distribution
of the fields over some geometric surface of the second order covering the
sources of the fields. The distribution of the fields of real sources is
presented in the form of the sum of electromagneCic fields of dipoles arbi-
trarily oriented and arranged within fihe above-menCioned surface. The di.-
poles used in modeling
D,~h~c~~~t~, ron),
where i=1, 2, 6 is the number of dipoles, are defined by six parameters: -
the components of the moment~li~l~(t) of the dipole along the axis of the co- -
ordinates and a~~l~ coordinates of the center of the dipole. The moment
- of the dipole M~1 (t) is the time function and in quasi-stationary appxoxi-
mation can be written in the form of
- M~t~(~) - N1~m~ e:co [1 ~wt - a~)) ~
where M~l~m is the dipole moment modulus; W is the frequency of the electro-
magnetic field; 4~ is the coefficient; and ~ is the r_oordinate.
For a rotating field: u= ~ E_ ~F+ for a traveling field: r= ,
e= x; and for a pulsating field: a= 0. Here, ~ is the pole pitch. _
In order to determine the parameters of the dipoles of an equivalent model,
it is necessary to have information about the arrangement of the sources and
their geometry. Therefore, in their analysis it is assumed that the
108 ~
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arrangement of the sources and their form are knewn and,consequently, the
� approximate arrangement of the dipoles is also known. In addition, it is
also assumed that the frequency spectrum of a real electromagnetic field does
not exceed 5000 Hz, which makes it possible to use the quasi-stationary ap-
proximation in solving a system ..f Maxwellian equations.
This article gives a method of numerical determination of the parameters of
specially oriented dipoles with whose aid it is possible to replace a real
field of a source at any distance from its su.rface.
1. The problem of the determination of the parameters of dipoles approximat-
ing a prescribed field is formulated in the following way. Let us assume
tha~ normal electric or magnetic intensities of the electromagnetic field
Fn~k are known at points Qk(k=1, 2, N) which are evenly distributed
- over a second-order surface S(of a sphere, circular cylinder, ellipsoid,
etc),c~ n~k are the corresponding normal intensities of electromagnetic
fields of dipoles which approximate the external field of the source. The -
problem is solved if the parameters of the dipoles are determined in such a
way that the error when the field of the source is replaced by the field
of the model,
N
- b = ~ ~Fn 1Q~ ~ ~n ~'K~~k ~ 1~
- k =1
is smaller than the prescribed number E(S 1� .
l L Q k
Q
If Fn(Q)~k are known, the system of equations (4) makes it ~iossible to deter-
mine the parameters of each of the G dipoles. To do this, it is necessary
- to solve a system of equations for N~ 6G. Due to the fact that the equa-
tions of the system are nonlinear, the most effective method is that of dif-
ferentiation with respect to the parameter [3,4]. The latter can be consid-
ered as an approximate method of solving a system of nonlinear equations.
2. The determination of the parameters of dipoles of a model of an electro-
magnetic field of a source is carried out in the following sequence. First,
we find the parameters of the first dipole satisfying normal intensities of
the field on the surface S at a certain number of points Qk. As a rule, a
dipole creating maximum known field intensities on S is selected. On the
same surface, normal field intensities created b the dipole at points ~k~
`k n~4)+k are calculated, and then the error ~~1~ and the remanent field with
intensity F~1)n(Q)Ik are calculated. Then, the intensities of the remanent
field are used to determine parameters of the second dipole of the model,
the error ~~2~ and the second remanent field with intensity F~2)n(4)Ik�
If necessary, the parameters of the third dipole, the value of the remanent
field and ~~3~ are calculated. Calculations are continued until the error
b~1) becomes less than E(S~t~ ~ E). ,
Thus, the problem of the replacement of a real external field of a source by
a field of a system of dipoles can be solved if the investigator has the neces-
sary information abaut intensities Fn(Q)~k of the electromagnetic field at
points Qk on one of the surfaces of the second order. When approximating
external fields of working electrical equipment, information about norcnaY
intensities Fn(Q)~k can be obtained in the process of the experiment.
110
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IR
1 1 '3 I4 ~S a~~
~ o~,
0
- d0~ - - - - - ~
~I ~P
I ,
_ - I
~ Figure 1
An example of the approximation of the magnetic field of an asynchronous
motor is given below. _
3. For a fixed moment of time, normal intensities of the field at points
Qk evenly distributed over the surface are measured on the body of the motor
or in the immediate vicinity of it and points with the maximum intensity are _
determined. It is assumed that the direction of the axis of the dipole and
its position are determined by the line connecting the points with the maxi-
= mum intensity, and the modulus of its moment M~l~m is faund by solving only
two equations from (4). Having determined the parameters of the first dipole,
we find that intensities created by it at points Qk on S, and the first re-
manent field, where points with the maximum intensity are also isolated, and
by them we find the second dipole M~2)m and the second remanent field. The
- process of approximation continues un~il the required accuracy is reached.
' Example. In order to construct a dipole model of an asynchronous squirrel-
_ cage moCor of the type A-42-4 with the following principal data (delta con-
nection of stator phases):
PH=2.8 kW; ~;v=314 c-1; p=2; n-1420 rpm; Gi - 0.20 m; d~,22 m
normal intensities HR~ k of a variable magnetic field are measured at points
- Qk uniformly distributed over the surface of a circular cylinder (coordin-
ates R, cQ , z) through Q~, k~.05 m, ,L1 CP k=90 degrees (Figure 1) .
The table gives the results of ineasurements of intensities HR(Q)~k in A/m
made with the aid of equipm~ent of the firm Bryui' and K'yer while idling
(UXX=230 V, IXX S.0 A) and in the short-circuit mode (UK~ = 80 V, I~3 = 10.0
_ A) .
111
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~ Table
\ uk ~ XUACCTO{I XOJ{ ~2~ ~\Op07'KOG 3a:dWK~H11C
_ 0 1 2 3 4 5 I 1 2. 3 4 5
Yk~ I I! I I I I I I
~
~ 0 I 450 I?75 G~10 770 S00 14,0 21,0 22,0 25,0 12,5
90 320 7:0 680 700 312 15,0 20,5 23,0 22,5 13,0
lE0 G~kO 1200 1150 1320 780 13,0 2I,0 21,0 22,0 12,0
' 270 280 G30 I 770 700 350 12,5 22,5 23,0 20,5 10,0 _
Key: 1. Idling
2. Short circuit
Radial dipoles are constructed (with an error of up to 10 percent).
j~Thile idling
~i M~,i 0,32 A~i2, zio = 0,05 M, Rio = 0,01 M, 'Pio = I~�:
DZ ~11;,~'-~ - 0,32 AM', z~o =-0,05 at, R2a = 0,01 M, ~20 = 180';
, D3 ;11~n~ - 0,095 An~~, z30 = 0,04 x. Rao = 0,004 nt, ~~o = 90�;
D4 M,n 0,095 AM~, z~o 0,05 M, R,o = 0,004 M, ~~o = 90�.
In short-circuit mode
Dl 1L4;,ii~ = 0,009 An~z, z~o = 0,04 nt, Rlo = 0, ~lo = 180';
D2 M;,,2~ = 0.009 Ae~', zso 0.0~4 nt, Rso = 0, 4'so = 180�~
D3 A9;,,3 0, 0028 A:~t~ ~ z~o = 0, 04 M~ R~ = U, ~~o = 90 ;
pQ M;,~4~ = 0,0028 An~$~ z~o 0,04 nt~ Reo = ~eo = 90�.
Here, ~1f~~~, J~S;,=2~, l~t;n~, Mm4~ are understood to be the moduli of the dipole
moments. In the instances when, as a result of solving the problems, it is
required to know not so much the structure of the field as its maximum value,
the number of dipoles can be reduced to a minimum (one or two).
Conclusion
The proposed method for approxima.ting external electromagnetic fields with the
aid of fields of dipoles is suitable for using at any distance from the source
and makes it possible to simplify the analysis of external electromagnetic -
fields of electrical equipment by using analytical methods both in calculat-
- ing passive shielding shells, and active compensating systems.
112
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� Bibliography -
_ 1. Apollonskiy, S. M. "Dipole Representation of the External Magnetic Field
of an Induction Magnetohydrod_ynamic Pump," MAGNITNAYA GIDRODINAMIKA
[Magnetic Hydrodynamics], No 3, 1976.
2. Tozoni,0. V. "Metcd vtorichnykh istochnikov v elektrotekhniki" [Mettiod -
of Secondary Sources in Electrical Engineering], Moscow, Energiya, 1975.
3. Davidenko, D. E. "On a Method of Numerical Solution of a System of Non-
linear Equations," DOKL AN SSSR, Vol 88, 1953.
4. Davidenko, D. F. "On Approximate Solution of Systems of Nonlinear Equa-
tions," UI~AINSKIY MATEMATICHESKIY ZHURNAL [Ukrainian Mathematical Jour-
nal], Vol 5, 1953.
COPYRIGHT: "Izvestiya vuzov SSSR - Energetika", 1979
10, 233
CSO: 1860
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Electromagnetic Wave Propagation; Ionosphere,
Troposphere; Electrodynamics
f
. UDC 621.372 ~
ON THE ELECTROMAGNETIC FIEZD IN THE VICIDTITY OF THE EDGE OF A CONDUCTING
HALF-PLANE
Moscow RADIOTEKHNIKA in Russian Vol 34, No 7, 1979 pp 66-69
[Article by Veselov, G. I.; Platonov, N. I.; and Acheyev, V. Ye., submitted
9 Jan 79]
[Text] Analyses of electrodynamic characteristics of a number of microwave
devices are often done on models containing, apart from smooth boundary sur-
faces, the so-called "geometric singularities", for example, sharp edges.
It is very important to have an accurate description of the electromagnetic
field (EMP) in the vicinity of the points of geometric singularity, parti-
cularly for developing effective algorithms for the calculation of complex
electrodynamic structures [1,2]. Analyzed below is the behavior of the field
in the vicinity of the edge of an ideally conducting half-plane for its dif-
ferent orientation in relation to the flat interface of inedia with different
relative permeabilities ~ 1, �l, and ~2, }~2 (Figure la)1.
fP= S~r
lo=SO? E>>f~r
E/,~f C2i~Y p
~2~f
2 1'
~ ~p ~_0 ~ ~ ~p=0
�nf
~ t3~{~.r SP-S~s -
~=~2-~r''~ Q~ ~-~2 61,
Figure 1
1. This work was prepared in connection with the discussion regarding the re-
port of A. A. Kirilenko et al "Analysis and Optimization of the Parameters
of Polarization Converters in the Form of Waveguide-Type Arrays with cones
for Scanning Antenna Systems" at the meeting of the All-Union Seminar on
Scientific Methods of Higher Schools on Applied Electrodynamics held on
16 Feb 1978 at MEI [Moscow Power Engineering Institute].
114
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Investigation of the EMP in the vicinity of the point of geometric singu-
larity can be done by means of the Meixner method [3] which is based on the
. expansion of the field components into generalized power serie~ with their
subsequent substitution in Maxwell's equations and imposition of appropriate
boundary conditions. In [4], analysis of the field is carried out in the
vicinity of the edge of an ideallq conducting wedge surrounded by three sec-
torial regions characterized by differer.t scalar permeability (Figure lb), '
Howeve?-, the field was not studied for a number of configurations occurring
frequently in solving applied problems. For example, no analysis wa~ done
- of a geometrical model shown in Figure la which is a concrete modification
of tlie generalized model . f Figure lb for W2 n, = 2n, e, = e,, �a = F+i�
(axis z is normal to the plane of the drawing of Figure 1 and coincides with
the edge of the wedge). -
In order to analyze the EMP in the vici.nity of the edge, we make up a super-
position of two solutions:
Ns, Ez s E~ = 0(rl+`); H~ - p(r-1+`) for r-~ 0, where p ~
is the smallest positive root of the equation
�')cosnt-0.
F�(T) ~ - l~+~~~ cos(2~,-r.)t-(1 -r (1)
The solution is characterized by the presEnce of singularities in the vicinity
of the edge of an ideally conducting half-plane only in the transverse com-
ponents of the magnetic field Ht,
2) Es~ N: = O(rt); H ~ O(r~+`), Et m p(r'1+`j foY r--s 0.
Here, ~"G is the lowest positive root oi' the equation2
/ e /
p~ (T) s( 1- E' ) cos (2~, 2-~- cos r.t s 0. ( 2~
~ ~
In this case, the singularities are in the transfers components of the electric
field Et.
In addition to these two solutions, in the general case, there is a possibil-
ity of another solution which was not taken into consideration in the analysis
of fields in [4] (see Table 1,1):
3) Er~ Ht - O~~); Ez � ~~r'): HZ m C. O(r') fOx' r-> 0.
2�There is a mistake in the formula (1.3.14c) in work [4] for the general -
case. The correct exgression for F~ (2) has the form
F~ (t) ~ ( 1 - E' J I cos i cOS ~4i -~P~) T - sin s sin ('P, - ~Ps )'~J -
~ E~ l ~a
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As can be seen, tlie field in this case does not have any singularities, how-
ever, it is important to note that the longitudinal component of the ma.gnetic
iield in the ~~icinity of the edge is a nonzero constant, which was pointed
out in [5].
~[n tlic~ j;eneir..il cri~:r, the field Ln Che vici.nity of tl~c pc~int of ~;eomeCric si.n-
t;i~l 3) combination interference of type (1, 1,...1)
caill certainly be present.
With an even number of signals (n > 4) combination interference of type
(1, 1, 1, 2) will certainly be pre~ent. _
150
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~
p ~l,>, f,2), (1, /,2,3)
~9 (l~2~22 _
~ _
1~
2 ~ R~~;~, 1,3) QB /1 f~
Q7 Q~ (1,1, l, 4~ 1
Q6
0,6 ~Z~3~ f , � _ ti,
U5 -y~ ~ ;~l ~3.~ ~ D,5 ~ { ~
O,y (3,4 C19 ~
~ ~ ~ I ~ `
~2,y ~z,~l 31~ 5 ~ o,z -
a~ js ~~~41 ~16~
q~ o> ` - -
p (i�),~~a,z) ~ P ~ h,�,,.il,~~r,>,>,31,~�,>,2,zJ,~~,,,�>z~
>,0 , >,D
0,9 (1,3.3 ~ ~ 09;
08 ,,9 i(2 ~ 2f ~ ~ aB ~ R�5,6
U7 (>,/,3f 1 ~ , a~ ~
' ~ s
_ ~ ~2,2�~) ~ ~ !�>yl'~- a5
~4 ~~z,~J (�z~ ~~~.s~ ~
Q4
43 ~ n ~3
Q2 ~ ; ~ I I
q~ ' q1
j~ 1,5 2 2,5 3 ~5 I('f ~ 1,5 z 2,5 3 .~5 Kf
Figure 1. Figure 2.
Taking into account the fact that ~or the ma.jority of nonlinear devices the
amplitudes of combination interference diminish w~th an increase in their
order of magnitude [6], it is possible to draw the following conclusions:
With a freq~.:~~.~y~ overlap factor of less than an octave and a random number
- of signals, the most dangerous is combitiation irit~rference of the third order
of magnitude of type (1, 1, 1) and (1, 2) ; with a frequency overlap factor of
greater than an octave, the most dangerous is combination interference of
the second order, of types (1, 1) and (2).
Therefore, fur the purpose of eliminating combination interference of lower
orders, it is advisable to select the frequency overlap factor at less than
an octave.
The resul~s gotten and these recommendations can be utilized in calculating
the technical characteristics of devices (i.n particular, n-signal dynamic
microwave ranges) and for the purpose of optimizing their passband.
151-152
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~ibliography
1. Kotel'nikov, V.A. NAUCHO-TEKHNICHESKIY SBORNTK, LEIS, No 14, 1936.
2. Antonov, O.Ye. and Ponkratov, V.S. RADIOTEKHNIKA, Vol 18, No 11, 1963.
3. Shaft, P.D. "IEEE Int. Conv. Record," Part 2, Nos 22-25, 1965.
_ 4. Sverkunov, Yu.D. RADIOTEKHNIKA, Vol 27, No 8, 1972.
5. Nikolayev, Ye.N. RADIOTEKHNIKA, Vol 28, No 2, 1973.
6. Sverkunov, Yu.D. RADIOTEKHNIKA, Vol 29, No 9, 1974.
OPYRIGHT� RADIOTEKHNIKA, 1979
255-8831 ~
8831
, CSO: 1860
153
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~ Microelectronics _
UDC 621.3.049.77.002.002.5
CHEMICAL INDUSTRY EQUIPMENT IN MICROEL~CTRONICS MANUFACTURE
Moscow KHIMIKO-TEKHOLOGICHESKAYA APPARATURA MIKROELEKTRONIKI (Chemical
Industrial Process Equipment in Microelectronics) in Russian 1979 signed
to press 24 Jan 79 pp 2, 311-312
[Annotation and table of contents from book by Aleksey Tikhonovich Myagkov
and Yevgeniy Mikhaylovich Korsetov, Energiya, 3,500 copies, 312 pages]
[Text] This book discusses designs, basic specifications and method of
engineering and designing of equipment for monitoring and controlling -
chemical industrial processes, equipment for dispersion and atmization of
aggressive media, equipment for preparation and batching of an aggressive
liquid and piping systems.
This book is intended for engineers and technicians working in the area of
design and operation of chemical process equipment in microelectronics
mant~facture.
Contents Page
Preface 3
Chapter One. Chemical Industrial Processes in the Manufacture of
Semiconductor Integrated Circuits ~
1.1. Chemical Industrial Processes and Types of Contaminants in
the Manufacture of Semiconductor Integrated Circuits 6
1.2. Methods of Cleaning the Surface of Silicon Wafers 8
1.3. Demands on Materials Employed in the Manufacture of Chemical
Processing Units 10 _
1.4. Influence of Operation and Design Factors on Corrosion of
Chemical Processing Units 13 F
1.5. Structural Diagram of Chemical Industrial Equipment 15
Chapter ~tao. Aggressive-Resistant Materials ~8
2.1. Basic Requirements on,Materials and Mechanism of Action of
Reagents on Them 18
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2,2. Basic Properties of Aggressive-Resistant Materials 22
~ 2.3. Interaction of Employed Materials With Chemically Acr_ive
- Reagents 38
2.4. Protective Coatings in Working With an Aggressive Medium 46
2.5. Methods of Computation oE Nonmetallic Components 49
Chapter Ttiree. Electromagnetic Valves 5~
3.1. Function and Types of Electromagnetic Valves 55
3.2. Computation and Design af Electromagnetic Valves 68
3.3. Sequence of Computation of Tractive Force of an Electro-
ma~netic Valve Drive 8~
3.4. Features of Operation of Electromagnetic Valves 97
Chapter Four. Pneumatic-Drive Valves 98
4.1. Function and Designs of Pneumatic-Drive Valves 98
4.2. Designs of Pneumatic-Drive Shutoff Valve Pairs 104
4.3. Valve Power Computation 109
4.4. Sealing Auxiliary Valve Components 117
Chapter Five. Pumps for Feeding Aggressive Media 124
5.1. Screw-Slot Pumps 124
5.2. Labyrinth Pumps 126
5.3. Bellows Pumps 139
5.4. Diaphragm Pumps 144
5.5. Computation of Intake and Delivery Valves 149
Chapter Six. Aggressive Medium Heaters 155
6.1. Theoretical Principles of Heat Transfer 155 _
6.2. Heating Sources in Aggressive Medium Heaters 163
6.3. Heat-Transfer Agents in Heat-Exchange Heaters 167
6.4. Design Features of Heaters When Working With an Aggressive
Medium 169
6.5. Heater Computation and Design 177
6.6. Heaters Employed in Chemical Industrial Process Equipment 187
Chapter Seven. Centrifuges 193 _
7.1. Principal Types of Centrifuges and Their Function 193
7.2. Processes Performed in Centrifuges 19$
7.3. Centrifuges Employed for Manufacture of Semiconductor
Integrated Circuits 201
7.4. Centrifuge Calculations 206
155
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' Chapter ~ight. Filters for Purifying Liquids and Gases 210
R,1. Pri.ncipal Types of Filters and Analysiti of Their Operation Zl~~
8.2. I~ilter.ing Materials 2L~
8. 3. 1~ llt4~r nesigns 231
8.4. Filter Calculations
Chapter Nine. Atomizer Nozzles 235
9.1. Theoretical Principles of Atomizing a Liquid in a Gas 240
9.2. Function, Types and Designs of Spray Nozzles 245
9.3. Calculation of Mechanical Atomizer Nozzles 24~
9.4. Calculation of Pneumatic Atomizer Nozzles
~ Chapter Ten. Aggressive Medium Sensing Elements 254
10.1. Function and Basic Characteristics of Sensing Elements 255
10.2. Level Sensors 259
10.3. Aggressive Medium Flow Sensors 265
10.4. Temperature Sensors 269
10.5. Pressure Sensors
10.6. Basic Characteristics of Sensing Elements Determined by 2~5
Operating Conditions
Chapter ~lPVen. Control Switches 280
11.1. Types and Function of Control Switches in Control Systems 28~
11.2. Designs and Operating Princi~~le of Control Switches
11.3. Calculation and Designing of Basic Components of Control 289
Switches 293
11.4. Command Devices of C~~n~rol Switches
Chapter Twelve. Pipe Fittings 293
12.1. Nipple Pipe Fittings 294
12.2. Flange Pipe Fittings 295
297
Appendices
Bibliography 305
[263-3024]
COPYRIGHT: Izdatel'stvo "Energiya", 1979
3024
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Photoelectric
UDC 621.382.2/.5
MEASUREMENT COMPONENTS AND PHOTOMULTIPLIERS _
IZMERITEL'NAYA TEKHNIKA in Russian No 6, 1979 pp ?1-73
[Article by L. I. Andreyeva, S. A. Kaydalov and B. M. Stepanov~
[Text] Domestic industry now produces over 200 types of photomultipliers and
photoelectric cells, forming a foundation for many devices and systems used in
optical and physical measurements in the many fields of science and technology.
The physical principle underlying the operation of photoelectric cells and
photomultipliers is the external photoeffect and secondary electron emission2
_ (time-delay less than i~'13 sec., thermal agitation noise 10"12-10'17 A. cm,
- photoelectric current gain up to i~81; it makes it possible to design photoelectric
cells and photomultipliers for measu~ ~~~~ent purposes that have subnanosecond
time resolution and high sensitivity to discrete photons [1-6].
Photoelectric cells and photomultipliers are widely used to measurement tech-
n~logy as the primary measurement transducers and, to a significant degree,
determining net measurement error. Metrologic characteristics of photoelectric
- cells, however, (1-3, 6-9 in Fig. 1) and photomultipliers (3-10 in Fig. 2) are not
calibrated [7-10] and only the circuit diagram of their parameters is indicated
[11,121. Questions of inetrologic support of photoelectric cells and photo-
multipliers are considered below based on R&D experience in measurement -
technology of photoelectric cells used in photoelectric colorimeters (FEK) and
photomultipi.iers used in cathode ray devices (ELU) [1-61. This problem is being
tackled with unified methods and on a legislative basis by the promulgation of
- a state system of guarantee of ineasurement unity [141, positions of inetrologic =
guarantee of the national economy [13] and creation of a unified system of state
standards for power engineering photometry [15].
The creation of ineasurement photoelectric cells and photomultipliers presumes
their devclopment as means of ineasurement in conformity with test circuits and'
systems of the state standards in the form of a composite set of devices based
on the principles of funetional, design, metrologic compatibility and inter-
changeability in terms of the conditions and features of operation.
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I~OIt U1~I~ICIi1L USI: ONI,Y
To r~llow for requir~ments, measurement photoelectric cells used in photoelectric
colorimeters (FEK) and electron photomultipliers used in cathode ray devices
(ELU) [1-61 hav~~ bc~en developed as a constituent part of the set of photoelectric
measurement '~ransducers for engineering photometry and measurement of the
parameters of high-speed processes in a spectral quantum energy emission range
from U.9 eV to l.5 MeV, dynamic range of over 200 dB, with time resolution of
- up to 3 x lU-1 ~ sc~conds. For interface with interactive links of ineasurement
devices, the photoelectric cells of photoelectric colorimeters (FEK) and
photomultipliers of cathode ray devices (ELU) have broad-band coaxial outputs
- with wave impedance of 50 and 75 ohms, and are provided with standard
connectors and cables of types SR and RK [16] (1-5 in Fig. 3). To match the
characteristics of ineasured emission fluxes and to assure the desired range of
photoelectric conversion at the input to photoelectric cells and photomultipliers,
standardized components are utilized: diaphragms, telescopin~ attachments,
diffusers, selective and non-selective filters, lenses, photometric spheres and
other components. According to operating conditions, photoelectric cells and _
photomultipliers may be used with or without auxiliary screened housings (6-11
in Fig. 3). ,
To guarantee measurement in an expanded spectral range, photoelectric cells
and photomultipliers are built with input windows and photocathodes whose
characteristics are cited in the table.
Measurement of parameters of powerful emission fluxes ranging from 10 to 102
watts is supported using photoelectric cells and additional attenuators. To raise
their time resolution or expand their dynamic range, photoelectric cells used in
photoelectric colorimeters (FEK) have increased field strength in the photo-
cathode up to 1-10 kV/mm; the anode and cathode are made in the form of
broad-band strip and ceaxial lines with wave impedance of 50 and 75 ohms,
anode assembly inductance is reduced to 10-9 H; and photocathodes have ,
resistivity no greater than one ohm per square centimeter.
To reduce the effects of dielectric design component charging, especially the
input window, screen electrodes and grids are employed. Parameters such as
photocathode surface, anode-cathode distance, anode voltage distinguish the five
type-sizes of photoelectric cells, assuring an optimum relationship between
required current and tolerable stray capacitance. To reduce dark currents,
increase the threshold of sensitivity and lengthen service life, the design of
photoelectric cells envisages inereased electrical insulation of electrode leads,
assuring leakage current no greater than 10-9 nA at voltages measured from 1
to 10 kV; this is done by using methods of oil-free evacuation to 10-9 mm Hg
and high vacuum getters. Mechanical and electrochemical treatment of high- ~
voltage parts is carried out to the lOth class of purity. To stabilize parameters,
photoelectric cells are technologically aged at increased gating voltages.
Measurements of parameters of average power emission fluxes in the range from
10-2 to 103 watts are assured by a group of photoelectric cells with increased
sensitivity (FEK-PCh)(5 in Fig. 1) and photomultipliers with fewer dynodes. In
the photoelectric cells FEK-PCh, photocathodes with negative electron medium
based on Ga~s and semiconductor silicon structures with gain of up to 103 are
used. In the SELU-F type photomultiplier (4 in Figure 1) and 32ELU-F (8 in Fig.
'
; 158
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2), electron multipliers with 1-5 stages and gain of 10 to 103 are used, while in
_ the 36EU-F type photomultiplier (9 in Fig. 2), microchannel MKP multipliers
with d~rk current of 2 x 10-11 A and gain of 2.5 x 104 are employed.
.
~ . ~
. ~
? ~ . _
,
; ~ ~
_
.
~ .
. ,
, , a ao _
9
. . ~ .
; _ , : _
Z J ~ g ; ? r
~ , ;
1 '6. 7 ~B 9~,, ID 1` Z 'J 4 5 6 i
Figure 1 Figure 2
. . . _,~...m.. _ _ ;
8. . i7 B 9,. : 9~ . 11 1
~Y~ ~
~ i:,' ,
;
~ : `
;
,
,
. , . . . : , i 7 : 3 . 5 ~
Figure 3
To measure weak emission fluxes in the range from 10-3 to 10-15 W on the basis
of electron multipliers of cathode ray devices (ELU) [1], type ELU-F type
photomultipliers have been developed. In terms of funetions performed and
design features, we can point oU.~photomultipliers for measuring parameters of -
maximally weak emission fluxes, high-speed photomulti~liers for pulsed photo-
metry, photoelectric cells with log conversion and variable photomultipliers.
Photomultipliers (FEU) for measurement of maximally weak emission fluxes have
several design technology features. Electron-optical systems of input chambers
have precision focusing and provide selection of useful signal photoelectrons with
a collection coefficients of at least 0.9. The first emitter has an increased
159
R(lU ~1FL'Tf'~TA T TTC~ l1T~TT.V
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coefficient of secondary emission-at least 10-30-wh'le the overall gain of
multiple stage electron multipliers comprises 106 to 10~. To reduce the effect
of such noise components as leakage current, gas-discharge processes, auto-
clectron emission, ion and optical feedback,special mounting insulators, retnining
rings, electrode screens, absorbent opticAl coatings and high-vacuum getters ~re ~
used. The manufacture of the photocathode by the method of vacuum
manipulator preclu~es the formation on the electrodes of the input chamber and
insulators of photomultipliers of photosensitive and emission-active films. In
these devices, the primary noises are induced by thermoelectronic emission,
- permitting measurements to be made at the level of photon noise of emission
flux under conditions of pulse counting. Inherent photomultipliers noise
~ comprises 0.1 to 103 pulse/second, according to the type of photocathode and
operating temperature.
1 2 3 a 4 yecrexTe:texocn
CnenTpaabxdtl AxanaaoE~, BxoAxce KaroA I ! pl I t(0,53), I i(I,OG). I ~Ce.
MKM O%HO T~ (\m). g /~t 3 9
9n)~or 5 A/Dr S A/sr 6~ A/IIr 6 A/nw
-3
I,2�10 5-8,1�10 ~ Al; Fe CuAIMgO 1,5�10 - - - "
CuBeO -3
~,1�10 5-8,1�10 7 Ba CuA1Mq0 1,5�10 - - - -
0,11-0.22 MgF= Cui: Csl 0,1; 0, 2 - _
0,22-0,8 YT-49 Ne,K(Cs)Sb 0,25 O,t 0,08 100
0,11-0, 9 MgF, GeAs(Cs) - - - _ 600
0,38-0,68 C52-1 Cs Sb(O) 0, 15 0,05 0.03 80
0.38-0.8 C52-1 Ne,1~lCs)Sb O, R5 0. 1 0.08 - I00
0, 22-1, 2 YT-/9 AgOCs (3-8) � 10 3(2-5) � 10 3(0.8-4) � 10~~ (2-5) � 10 ~ 30
0,38-1,3 C52-1 AgOCs (3-8) � 10 3(2-5) � 10 3(0.8-=) � i~-3 ~~-5) � 10 ~ I 30
[Key: 1. spectral range, microns; 2. input window; 3. cathode; 4. sensitivity; 5.
electron/photon; 6. A/W; 7. A/lm.]
The 31EI~U-F device (10 in Fig. 2) was developed for pulsed photometry. A
distinctive feature of its design is the placement of the frame photocathode on
a disc cold-conductor connected by a metal- lass weld with a thermoelectric
cooler; this reduces its dark currents to 10 ~ A at gains of 107 to 108 and
retains the advantages of high-speed photomultipliers [4].
tn high-speed photomultipliers, the electron-optical systems of the cathode
chAmbers and secondary-electron emitters provide an electron dispersion of at
least 0.5 ns, input lead inductance does not exceed 10-8 H and the circuits of
the final emitters contain compensating capacitors af 10-9 to 10-8 farads
capacitance. The output signal of the electronic multiplier is made inthe form
of a screened broad-band coaxial-strip line with wave impedance of 50 and 75
ohms. To increase sensitivity and reduce noise currents, the photocathodes of
high-speed photomultipliers of the 18ELU-F type (7 in Fig. 2) are manufactured
with the vacuum manipulator.
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To measure in the dynamic range 107 to 10g based on high-speed devices,
photomultipliers are manufactured with log conversion, ELU-FL (6 in Fig. 2) and -
variable photomultipliers , ELU-FU and ELU-FS. Log conversion is done by
li~niting the current by a space charge in the intermediate or output stages of -
t~e secondary-electron multiplier by selecting the distribution of potentials on
the electrodes.
In variable photomultipliers, a system of electrostatic control of photocurrent
density with minimum potential is placed between the photocathode and the
input stage of the electron multiplier; it is controlled by distribution on elec-
trodes of the cathode chamber. Variable photomultip liers have transconductance
of up to 1 A/V, suppression coefficient of 104 to 105 and working frequency band
of up to 1.5 GHz.
In conformity with [14], metrologic characteristics of ineasurement devices are
characteristics exerting an effect on results and measurement error. Cer-
tification of inetrologic characteristics of photoelectric cells and photo-
multipliers lets us measure characteristics of test processes with desired error;
to compare and select photoelectric cells and photomultipliers according to
known conditions of ineasurement and required accuracy; and to estimate error
_ of complex devices and sytems according to desired functions and structure.
The list of problems to be solved suggests how important it is to standardize the
set of inetrologic characteristics of photoelectric cells and photomultipliers as
measurement devices. Apparently the set of standardized metrologic cha-
racteristics on one hand should not be extremely great, while on the other hand
it should reflect the basic physical and technical properties of devices and be
expressed in form and units permitting simple assessment of ineasurement
results of parameters and their conformity to the standards. When photoelectric
cells and photomultipliers are used as the primary rneasurement transducers, the
measurement result will be affected by three groups of parameters. First of all,
emission parameters at the input, matching of input window and photocathode
with emission parameters. Measurement errors at the input may be caused, for
example, by a partial miss of the photocathode by emission, multiple reflections,
instability of three-dimensional arrangements and distribution of emission and
other random factors. Secondly, parameters supported b~ photoelectric cells or
photomultipliers themselves. Thirdly, parameters of processing and mea-
surement devices for electrical signals at the output of photoelectric cells and
photomultipliers, matching of impedance, amplitude and phase-frequency cha-
racteristic parameters.
From the viewpoint of creating and using measuring photoelectric cells and
_ photomultipliers, the second group of parameters is most important. Despite the
diversity of photoelectric cells and photomultipliers, it is possible to specify
general methods of mathematical description of their characteristics, con-
sidering the aspects of photoelectric conversion independently of the principle of
action and esign fetztures. In mathematics, the term "statement" corresponds to
the term "photoelectric conversion"; this establishes the relationship between the
elements of the two sets and correlates each element of one set to some
element of another set. From the mathematical viewpoint, photoelectric cells
and photomultipliers having emission flux at the input and an electrical output,
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are a realization of a photoelectric conversion statement, since a transient value
of output electrical signal corresponds to each transient value of input photon
flux. From the viewpoint of physical processes, spectral sensitivity
is the statement of photoelectric conversion; it appears in the equation
~ = ly.~~1,~ mg ~Of � \1~
where I is output current; (D is power emission flux striking the photocathode;
and Ip is noise current.
As was shown in studies [3, 151, only gradation and calibration of the
device in energy units guarantees the unity of ineasurements in the entire
spectral rang~ of electromagnetic emission. Measurement error, in conformity
with (1), may be represented in the form
em, e~, 1 ~o + ,!~t (2)
~3 t9 J ~
From (2), it follows that measurement photoelectric cells and photo-
multipliers should retain unchanged spectral sensitivity in the entire range of
measured quantities in order to have a high signal-to-noise ratio and support
adequate output currents with an error of Q, I. ~
As was shown in [3l, the basic metrologic chara~teristics of photoelectric ~
cells and photorrultipliers, when measuring parameters of emission pulses using
the ~ethodof high-speed oscillography, are connected by the relationship
r
~ h (t) dt ~3~
~3 = �e ~
TjT.i3 (7l)
- where T is pulse repetition rate; h(t) is beam deflection on the oscillograph
screen at time t; f is oscillation sensitivity; and Z is wave impedance of the
measurement tract.
In conformity with (3), amplitude of beam deflection on the oscilloscope
screen may be found by the formula
h(t) = IjZ -
- If d denotes the beam 0 on the screen, then it is possible to find the
relative error of amplitude measurements with output current I:
I = d/IjZ.
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On the other hand, relationship (3) does not explicitly take into account the
dynamic characteristics of photoelectric cells and photomultipliers. It is
~therefore necessary to take both amplitude and time measurements with error
t on the time axis,. whereas the standardized time resolution
Ot of photoelectric cells and photomultipliers [3, 4, 17) also determines the
accuracy of photoelectric conversion of emission pulses of desired width t N:
~tl 100 c dt. (4)
- As we know [17), time resolution is determined as a function of conditions of
; measurement of pulse characteristics, transient characteristics or the complex
coefficient of transmission set by the amplitude-frequency and phase-frequency
characteristics of the photoelectric cells and photomultiPliers. It should be
noted that according to classification in [14], photoelectric cells and photo-
multipliers can be related to the first subg~oup of the first group of
measurement devices. Thus in addition to the basic error components already
considered, it is sometimes necessary to standardize funetions of effect caused
by the possible deviation of inetrologic characterist~es from rated values and
determine additional error under varied operating conditions.
For certification of inetrologic characteristics of photoelectric cells and
photomultipliers, as was indicated above, it is necessary to eliminate re-
production error of parameters of emission at the input and measurement error
at the output. This condition is satisfied using special meter devices of the
appropriate testing circuits of state references. The purposeful elaboration and
metrological certification of ineasurement photoelectric ~ells of photoelectric
colorimeters (FEK) and photomultipliers of cathode ray devices (ELU) now
assures the creation of new methods and means of optical-physical mea-
surements with an error of photoelectric conversion no greater than 15 percent.
References
1. Andreyeva, L.I. and Stepanov, B.M., PTE No 3, 1962.
2. Andreyeva, L.l. nnd Stepanov, B.M., PTE No 5, 1962.
3. Andreyevt~, L.I. and Stepanov, B.M., IZMERITEL'NAYA TEKHNIKA No 8,
1965.
4. Andreyev~, L.I. et al., in: Impul'snaya fotometriya [Pulsed Photometry]
Mnshinostroyeniye Publishers, 1972.
5. Andreyeva, L.I. et al., PTE No 3, 197Q.
6. Andreyeva, L.I. et al., PTE No 6, 1976.
GOS'I' 17485-77 "Photoelectric cells, electrical vacuum type. Basic
specifications".
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t'Utc ur r~~.iriL vo,~, v..,~.
8. GOST 19798-74 ~~otoelectric cells for general-purpose devices. General
technical specifications".
9. GOST 20256-75 "Multipliers, photoelectronic, photoelectric cells. Terms,
= definitions". -
10. GOS'I' 17470-72 "Multipliers, photoelectronic. Basic specifications"
ll. GOS'C 21316.0-75 "Multipliers, photoelectronic". ~
12. GOST 11612.0-?5 through 11612.16-75 "Photoelectric cells. Methods of
measurement of parameters".
13. GOST 1.25-76 "GSS. Metrologic Support. Basic Assumptions".
14. GOST 8.009-72 "GSI. Standardized metrologic characteristics of inea- _
- surement devices"
15. Kotyuk, A.F. et al., IZMERITEL'NAYA TEKHNIKA No 3, 1976.
16. Spravochnik konstruktora radioelektronnoy apparatury [Handb~ok for Radio
Engineering Designers], G. R. Varlamov, editor, Moscow, "Sov. radio", 1973. .
17. Andreyeva, L.I. et al., in: Impul'snaya fotometriya, Leningrad, Mashino- _
- stroyeniye Publishers, 1975.
COPYRIGHT: Izdat~l'stvo s~andartov, 1979 ~
[249-8617] _
8617
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UDC 621.383.52 -
PHOTODIODES AND PHOTORESISTORS BASED ON GaSe -
Moscow RADIOTEKHNIKA I ELEKTRONIKA in Russian No 7 1979 pp 1430-1432
[Article by G. B. Abdullayev, N. B. Zaletayev, A. Z. Ma.medova, T. V.
Rudobol and V. I. Stafeyev: "Photodiodes and Yhotoresistors based on
GaSe"]
r.
[Text] The study concerns research on the photosensi-
tivity, noise and threshold characteristics of photodiodes
. and photoresistors based on monocrystals of gallium mono-
salenide.
The diodes were fabricated by fusing tin (the rectifying
- contact) at T= 500-600�C. Specimens of ~aSe of the p-
type with hole concentrations of 1015-101 cm"3 and mo
bility 30 cm2/v�sec were used.
It is shown that the diodes have photosensitivity in the
range of the spectrum~= 0.36-0.65 mkm and the long-wave
sensitivity threshold of diodes with an s-shaped volt-
ampere characteristic reached 1 mkm.
The detector property D* at the maximum spectral sansi-
_ tivity ~ = 0.6 mkm of diodes with str i ght s~ift
U= 10 vmand T= 300�K achieved D* = 101~ cm Hz2 vt-1,
which is a factor higher than with photoresistors of GaSe.
Among semiconductor compounds AIIIBIV gallium monoselenide has a number
of interesting properties: a significant photosensitivity to visible radia-
tion, photo- and electrolumine~cence in the presence of induced radiation
with rapid electron excitation and so on and, due to this, it may be used
to create sources and receivers of radiation.
Earlier we have reported on some of the electrical properties of diodes
based on gallium monoselenide (1).
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In this study the resul-s are given of research on the photosensitivity,
noi:e and threshold characteristics of diodes and photoresistors of gallium _
monoselenide at temperature intervals at 77-300�K.
For the base material of the diodes and resistors was used unalloyed and
tin-alloyed specimens of gallium monoselenide of the p-type obtained by the
Bridgeman method. The conce tration at~d mobility of holes in the mono-
crystals of GaSe was p= 101~-1016 cm 30 cm2/v�sec with 300�K.
Wafers of gallium monoselenide were obtained by shearing from slabs of
GaSe perpendicular to the direction of the "C" axis. As a result of this
- shearing a smooth, mirror surface was obtained which did not require further
processing.
The diodes were manufactured by fusing tin (the p-n junction) at a tempera-
ture oE 500-600�C and indium (the ohmic contact) at T= 200-300�K on _
different sides of the GaSe wafers. The specimens were illuminated on the
side of the rectifying contact. The surface area of the diodes was S^~0.5
- 1 ~n2.
' Used in manufacturing the photoresistors were specimens ~f GaSe and Ga5e
alloyed with tin (Sn) with the dimensions 3 X 3 X 0.2 mm at the end of
which the ohmic contacts were developed by fusion with indium. The
specific nonluminous resistance pT of the GaSe photoresistors was on the
order of 102-104 ohm�cm and of the photoresistors of GaSe(Sn)^~105-106
ohm� cm.
Measurements were made with Rspec 7~ Rload� The spectral characteristics
were measured with a monochromator DMR-4. The integral photosensitivity
was measured at a modulation frequency of 100 Hz and illumination of 100
lk from an incandescant light ("A" source). The photoresponse was re- -
corded on an SI-19B oscillograph. The noise characteristics were measured
with a selective micro-voltmeter with an effective band ~,f ^'6Hz. ~
I,,+a -
Sn 20
~ jraSe
1,5 �
In ~p
RS .
_ -40 -30 -ZO -10
~
- 10 ?0 3U U,e
-45_
Figure l. Nonluminous Volt-Ampere Characteristic of a
Photodiode of GaSe
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I~,OmN.ed _
f00 - jr ~3 1 ~a~ t
~
80 ~ xK ~ I
u0 99 "~.x 1
~
~ ~'~Z 1
. p k
'i ~ \
0 `
` Q35 0,4 0,45 0,5 Q55 0,6 MKM
Figure 2. Spectral Characteristics at T= 300� K
1, 3-- diode with forward and reverse bias;
2 photoresistor of GaSe
, ~1;"cnzu'hem'
~
- SI, MalnH: U~. nxQ SI, na/nx
10~ 102 k 5 3_~iJ'n�
D . I
~o� ~o' D* ~ 1 ~o~
~ 6 .
x ~
10~ 10~ ~ x~ x 10"? -
U,~ Z~~
SI
l0a 10~ ~U~ t
10'~ 10� 10~ U,C
Figure 3. Dependences S, D* and UW on Applied Voltage
for Diodes (1~3,5) and Resistors (2,4~6)
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U.~.�rxo (/~,ti .
~0? ~ .
2�~ '
1..',Sri
t01 ~ IG~
~o.se
. �
IO� ~ � , 10'~
1~' 10z 10~ f0~f,;~
Figure 4. Noise Spectra for a Diode (1, Y-axis to the left)
and a Photoresistor (2, Y-axis to the right) from
GaSe with T= 300� K
On the straight leg of the volt-ampere characteristic (VAKh) sections of
several diodes with negative differential resistance (ODS) of the s-type
were observed. The voltage of the junction in the section with ODS
amounted to 20-40 volts (fig. 1). Since the base of the diode was manu-
factured from high-ohm material the basic voltage gradient with forward
bias fell within the base and not at the p-n junction. In connection with
this, the voltage across the diode considerable exceeded the magnitude of
the potential barrier at the p-n junction. _
The diodes possessed photosensitivity in the range of the spectrum~ = 0.36- -
0.65 mkm (T = 300� K). The long-wave threshold of sensitivity of the speci-
mens with s-shaped VAKh reached a~- 1 mkm. The maximum spectral sensitivity
of the resistors and diodes with forward bias was found in the wave-
leng ~i range ~ m= 0.56-0.6 mkm (fig. 2, curves 1 and 2). With reverse
bias ~ m of the diodes was shifted to the short-wave range of the spectrum:
m= 0.44-0.5 mkm (fig. 2,curve 3).
The integral current sensitivity of the diodes and resistors Sl increased
linearly with the applied voltage (fig. 3, curves 1 and 2) and reached
5-102ma/lm in diodes with forward bias and 1-10 ma/lm in the resistors.
Thus, the diodes possessed higher sensitivity than the photoresistors of
the same material. .
The voltage of the noise U w in the ma.jority of diodes and photoresistors
at room temperature was also increased almost linearly with an increase
in applied bias (fig. 3, curves 3 and 4).
In all of the studies photoresistors and diodes in the frequency range
~ f=25 - 2�104 Hz noise with a spectrum of the type 1/f was observed
(fig. 4). When the temperature was lowered to 77� K the voltage of the _
noise was sharply reduced. The nature of the dependence of U u1 remained
unchanged here. The excess noise with a type 1/f spectrum in the photo-
resistors and diodes may be related to the process o� manufacturing the
- contacts and processing the surface of the specimens.
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The detector property at m~iximum of the spectral sensitiviiy ~1=1Q.6 mkm in
diodes with forward bias U= lOv and T= 300� K reached 10 1 10 ~ cm Hz~2vt-1
which~corresponds to the cut-off power P~ _(UW /RH )/S1~ f= 10'12-10-13
- vt/Hz~. The ~value of the detector property in the photoresistors was
approximately a factor lower. '
BIBLIOGRAPHY
1. Aliyeva, M. Kh.; Mamedova, P. F.; Mamedova, A. Z.; and Muradova, G. A.
DOKL. AN AzerSSR, 1972.
[272-8945]
COPYRIGHT: Izdatel'stvo "Nauka~" "Radiotekhnika i elektronika," 1979
8945
CSO: 1869
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~
Radars, Radio Navigation Aides, Direction Finding
UDC 621.396
EFFECTIVE ALGORITHMS OF MAXIMALLY PROBABLE PROCESSING OF RADAR SIGNALS
Moscow RADIOTEKHNIKA in Russian Vol 34, No 7, 1979 pp 8-14
[Article by V. I. Chaykovskiy, submitted after revision 8 Sep 1978]
[Text] In many instances, radar searching (simultaneous detection and mea-
surement of the parameters of useful signal) amounts to a maximally probable
evaluation of these parameters and the verification of the truth of the hy-
pothesis about the existence of the signal in the observed realization
Y(~) = s(t, a) X(t t E 7~. (1)
where s(t,[X) is the determined useful signal with a vector of unknown pa- _
rameters 0(; x(t) is the Gaussian noise with a known correlation function
[1]. The above procedures can be performed on the basis of the processing
of the complex envelope of the observed realization
Y(t) = S(t, -I- X(t), t E T (2)
as a result of the formation and extreme analysis of the probability function
or sufficient statistics of the vector of informational Farameters of the
signal component. The switching to the complex envelope of the realization
of the band-pass signal being studied is particularly advantageous in digital
processing systems. In this case, due to the low-frequency nature of the
spectrum o� the complex envelope, the quantification frequency and the aper~
ture error of information quantization during analog-digital conversion de-
crease. Moreover, the complex nature of the representation of information
is well in agreement with the functional characteristics of the procedures
of digital spectral analysis which are used widely in modern processing sys-
tems.
Let us represent the complex envelope of the observed realization in the form
of a column vector Y in a complex unitary space by the expression
Y =TSp T X, (3) .
where the complex amplitude 7= Ae~~ allows for the influence of the ran-
dom initial phase of the useful signal , the information parameters t�~
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are determined by the time shift of the signal 'L and the displacement of
its carrier frequencyl 1J, and the random vector X with a normal distribution
law and a known covariance matrix
A = E {XX''} ~4~
represents an additive interference. Here and hereafter, the line symbolizes _
the operation of complex conjugation. Having used a known expression for
~ a normal distribution law of a random vector in a real unitary space [3] and
having transformed it with consideration for the peculiarities of the scalar
product in a complex unitary space, it is possible to obtain an expression
for a probability function of a vector of information parameters ~ averaged
over the area of variations of the interfering parameter (random phase
-
L(~9, Y)=2~ Sexp~Re{Aexpi~?S~.1-~Y}Jd~=1o(A~Sp~1-IY~}, (5)
0
where Io is the Bessel function of the first kind of the zero order
whose argument is the biZinear Hermite form of the vector S~ of the infor-
mation part of the complex envelope of the useful signal and vector Y of the
~ complex envelope of observation. The matrix n-1 of the bilinear form
is formed by inversion of the covariance matrix (4), It is evident that the
bilinear Hermite form is a sufficient statistic of the vector of informational
~ parameters ~ and forms above the plane of its possible variance the surface
1(~, 1') = I ~~1,-1Y;, (6)
whose extreme analysis is aquivalent to the analysis of the likelihood func-
tion (5). Consequently, in the practice of radar searching Che maximally
probable processing (MPO) oriented toward detection of a useful signal and
the evaluation of its informational parameters can be reduced to
the formation and subsequent extreme analysis of the surface of sufficient
statistics j( Y). Thus, (6) is a matrix form of a basic algorithm of _
maximally probable processing of a complex envelope of a radar signal with
a random initial phase under the conditions of a Gaussian interference.
Having made use of the property of associativity of the multiplication of
matrices, algorithm (6) can be represented in the form of a modular value of
the scalar product of the vector of observation Y and a certain reference
vector GS
G~ S~A-1'
formed by the products of known matrices S~ and n-1*). Thus, the expres-
- sion
�In the general case, vector ~ can also allow for other parameters of a -
useful signal carrying useful information.
*Reference vector G~ can be considered as a solution of the system of linear
~ equations ,A G~= S ~ .
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1(~5, Y)=1(~~~ Y)~, (8)
representing each count of sufficient statistics in the form of a modulus
of the coefficient of mutual correlation or scalar product of the vector of ~
observation and a certain reference vector is true. Such an algorithm can
be called an algorithm of correlation of copies in the space of complex en-
v~lopes. A simplified functional diagram of the MPO system on the basis of
the correlation of copies is shown in Figu~e la, where symbol M designates
the procedure of the decermination of modular values, and symbol H designates
the procedure of the accumulation of pair products. As follows from (8),
the formation of each count of sufficient statistics l({3, Y) in an N-dimen-
sional space is materialized as a result of the fulfillment of N operations
of complex multiplication.
~ H y P. y n~ M B -
G~ Q G(kJ a h
y P y ~2~ ~ e
n~ N rr nf ax~ M
G~p+~~ C~~ ~d
Key: 1. PF Fourier transformation
2. OPF inverse Fourier transformation -
More economical MPO algorith~ with respect to the amount of computations can _
be obtained if we represent vector S~ as a sequence of counts. If a signal
with zero initial values of parameters and y has a complex envelope
S(t), then the complex envelope of the same signal with arbitrary values of
these parameters is, as is known [4], equal to
Sa (t) = exp tY (t - ~)l S -
Let us assume, as it is taken in the theory of discrete Fourier transforma-
tion (DPF) [S],that W is the symbol of the latticed complex exponential func-
( 2~ 1
tion explt N(�)~, and symbols r, k, q, and p respectively denote dimension-
less values of the displacement of frequency current time t, shift
and current frequency (~J . Then, the following representation of vector SR
in an N-dimensional complex-valued space is true:
$p - W- r(k-v?$ (k - 9), k E N.
The observation vector Y and interference vector X can be represented in an
analogous form: Y-1((k), X=X(k), kEN. In this case, the elementsof the
covariance matrix are defined by the expression
~kl = E{X (k) X(i)}, k, 1 E N
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and, due to the stationary state of the interference, possess the properties
of diagonal symmetry \~k! -~!k) and invariance to shift ~~`kt =~tktalc~+v)I�
The elements of the matrix .~"1 possess the same properties.
_ Having made use of the notations introduced above, let us represent the suf-
f icient statistics (6) in the form of
t~r~ Q+ Y~ = I w`� I�(~ w-'kS ~k - 9)~TA-~Y ~k) I, r, k E N.
By virtue of the above-mentioned property of invariance of the elements of
matrix /1"1, the follo~oing relation is true
1~r~ 9~ i') _ w-''~S ~k))T.~-~Y (k 9) I~ T� k E N~
which can be represented in the form of a modulus of scalar product
l(r~ 9~ 1t) = I~i W-`k~ ~k)~~ Y(k -I- 9)) I~ T+ k E N+ ~9)
where r,T (k) -ST (k)11-' is a certain reference vector coinciding with the
solution of the system of equations /~.G = S. By definition, a scalar pro-
duct can be written in the form of sum of pair products
l~r~ q� Y)=,~~(k)Y~k~- 4~W-~kl~ kEN� (10)
~ �
Consequently, the sufficient statistics coincides with the DPF modulus of the
sequence of pair products of elements of the reference vector and the sliding
vector of observation which, in technical applications, is called coXrelation
product
- 1(r, q~ 1f) _ ~ F(G (k) Y(k -I- 9)) I~ r, k E N.
(11)
In this expression, F=~ W-,kq is the DPF inatrix.
The MPO algorithm represented by expressions (10) or (11), with respect to
its informativeness, is equivalent to the algorithm of the correlation of
copies (8) and amounts to the determination of modular values of the sequence
of the DPF of correlation products of a fixed reference vector which is de-
veloping with time, and the vector of the complex envelope of observation
at each step of its sliding. The Fourier transformation of each correlation
product forms at points ~r a latticed copy of the cross section of the sur-
face of sufficient statistics ~((3, Y) parallel to the axis of the shifting
of frequency ~ with a fixed shift tq. Algorithm (11) which accomplishes the
MPO on the basis of spectral analysis of a correlation product can be called
correlation-spectral algorithm. A simplified functional scheme which real-
izes such an algorithm is shown in Figure lb, where symbol PF denotes the pro-
cedure of Fourier transformation, and the remaining notations coincide with
those adopted earlier.
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When the correlation-spectral algorithm is used in an N-dimensional space,
first there forms an N-dimensional correlation product G(k)Y(k~q), on which
N operations of complex multiplication are spent. Then the obtained sequence
is transformed by means of the procedure of fast Fourier transformati~n (BPF)
on w}~ich additional 0.5N log2N of multiplication operations are spent. Tt~us,
the total expenditure amounts to N+O.SN log2 N such operations. Since asa re-
sult of this, there form N counts of sufficient statistics corresponding
to various values of frequency shifts, then the proportionate expenditure of
operations of complex multiplication per one its count are equal to
N+ o, ~v log, N_ 1-;- 0,5 log2 N~ N
and always less than the corresponding expenditure when MPO is realized on
the basis of direct correlation of copies. If instead of the BPF procedure
the standard DPE procedure is used in calculations, the effectiveness of
the algorithm is lowered approximately to the level of the effectiveness of
the algorithm of direct correlation of copies.
Another economical MPO algorithm can be obtained by using expression (9).
- Since the operator of discrete Fourier transformation, by virtue of its or-
thogonality, is unitary, and, consequently, maintains the value of the scalar
product [6], then the following relation is true:
j~r~ 9, 1j) - I IF I W-`kG ~k)J+ F(Y (k 4)~)
In accordance with the known [5] DPF theorems, the first co-factor of scalar
product can be reproduced in the form of F(W-~"G(k)] =G(P-I-r) , where index
r symbolizes the shifting of the vector G(p)=FG(k) along the axis of fre-
quencies. By definition, Fourier transform of the 2nd co-factor coincides with
the sliding Fourier transformation (SPF) Yq(p) of the vector of observation
F IY J- 9)~ = Ya ~P)~ Pt k E N.
- Thus, the sufficient statistics is equal to
l(r, q, 1') = I~~ ~P r)� Ya ~P)i P E N ~ 12~
and MPO is reduced to the sliding Fourier transformation of the vector of
observation and determination of modular values of scalar products of the
vector of the sliding spectrum Yq(p) and spectral images G(p+r) of the
reference vector G(k) with prescribed frequency shifts r. The procedure of
the determination of scalar products is accomplished over the entire set of
possible frequency shifts vr at each step q of the sliding of the vector
of observation, As a result of this, there form latticed copies of sections
of the surface of sufficient statistir.s analogous to those obtained in the
realization of the carrelation-spectral algorithm (11). Algorithm (12) which
realizes MPO on the basis of SPF can he called spectral-correlational since,
unlike (11), spectral analysis is realized during the first stage of process-
ing, and the correlational product of spectral images is formed during the
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second stage. A simplified functional scheme of the system realizing tl:e
processing on the basis of SPF is shown in Figure lc, where the symbol P~'q
denotes the SPF procedure.
For the formation of each count of sufficient statistics, when SPF is real-
ized on the basis of effective recurrent procedures expending on each cycle
of sliding analysis N operations of complex multiplication [7], it is neces-
sary to perform only (1-FQ) such operations where Q is the active dimension-
ality of the spectral image G(p) of the reference vector G(k). When proces-
- sing radar signal with a small size of the base, this value is substantially
smaller than N and the spectral-correlation a?gorithm is not any less effec-
tive than the correlation-spectral algorithm. The algorithm is particularly -
effective in processing simple audio signals (Q=1), In this case, the ref- `
erence vector G(p) degenerates into a one-dimensional vector, which elimin-
ates the necessity of the formation and summation of the elements of the
correlation product in the spectral region. Thus, the structure of the pro- ~
cessing system issimplified as a result of the elimination of the complex
multiplication device and digital accumulator, and the number of complex
multiplications spent on the formation of each count of.s,ufficient statis-
tics decreases to one.
The algorithms (11) and (12) examined ~ibove make it possible to form latticed
copies of cross sections o: the surfac~~ of s~fficient statistics parallel to
the axis of frequency shifting. Havin;; used (11), it is possible to obtain
the effective MPO algorithm forming cross sections parallel to the time-delay
axis. As can be shown [5], the Fo+~~ier transform of the vector [G(k)xY(k#q)]
is equal to the convolution of the co-factor spectra. Since -
FG (k) = a~P), FY (k 9) = W P4Y ~P)~
then
l ~r~ ~1 ~ Y~ = I ~ G ~P - r) Y ~1~) W v4 I = ~ F
(p (p _ y
~P)I I� ~ ~3 ~
D
Thus, the sufficient statistics is defined as the modulus of the inverse
~~urier transformation o� correlation product of the reference vector G(k)
and the vector of observation Y(k) in the spectral basis with all possibYe
values of frequency displacement ~ r and shift tq. The algorithm of MPO -
corresponding to such definition of sufficient statistics in the first stage
amounts to the determination of the Fourier transformation of Y(p) of the ob-
servation vector and the formation of correlation products [G(p-r)Y(p)] by
the set of the varying parameter of frequency shift r. The spectral image
G(p) of the vector of the reference signal in this case is assumed to be
known. The second stage of processing amounts to inverse Fourier Transfor-
mation of the formed correlation products. Each such transformation, after
the determination of modular values, forms a cross section of the surface of
sufficient statistics parallel to the axis of shifts q in entire interval of
the determination of the vector of observation Y. Since the above algorithm
realizes the Fourier transformation both in the first and second stages, it
can be called spectral algorithm. A simplified functional scheme realizing
such an algorithm is shown in Figure ld where symbol OPF denotes the procedure
of the inverse Fourier transformation.
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In order to eliminate boundary distortions of the results of the processing
on the basis ot the spectral algorithm, the dimensionality M of the vector of
observation must correspond to the full range of possib le delays of tlie use-
fu1 signal and, consequently, to be substantially greater than its dimension-
ality N. This lowers the effectiveness of the algorithm. Sectionalization
of input information and its processing in mutually intersecting intervals
remove the above-mentioned defect. Organization of sectionalized processing
full,y coincides with the known processing [8] oriented toward obtaining ape-
riodic convolution in a space of real samples and does not require additional
explanations. The dimensionality of the vector of observation in sectional-
ized processing by the spectral algorithm usually is taken to be equal to 2N,
so that for the formation of one count of sufficient statistics (13) with the
use of the procedures of the fast Fourier transformation, approximately
(log: 2N ~ operations of complex multiplication are used, where Q is the -
active dimensionality of the spectral image of the reference vector G(p). -
Thus, the spectral algorithm is somewhat less effective than the correlation-
spectral algorithm, however, it always remains more economical than the algo-
rithm of direct correlation of copies.
The four MPO algorithms for complex enve~opes of radar signals are well suited
for realization on the basis of the use of digitnl computers, since, all other
things being equal, they minimize the ape-ture error of quantization and quan-
tification frequency at the stage of analog-digital conversion. Three of the
four algorithms make use in one or another form of spectral representations
of information, which makes it pos~ible, as a result of applying effective
procedures of the fast Fourier transformation or recurrence analysis of the
sliding spectrum, to reduce the volume of computations in comparison with that
for the realization of the traditional algorithm of the correlation of copies.
Moreover, the complex-valued nature of the input information agrees best with
the functional peculiarities of various procedures of the Fourier transforma-
tion, which additionally lowers the computation and equipment expenses. _
Since, as follows from the formulas representing effective algcri*_nms, in the
absence of an interference, the surface of sufficient statistics 1(r, q, S)
coincides with the surface of the body of mutual ambiguity of useful sigr~als
(Woodworth [4]}, such algorithms should be considered as algorithms of infor-
mationally complete representation of radar information. The completeness
condition is always satisfied when the steps of quantification with respect
to time and frequency are selected in accordance with the requirements of
Kotel'nikov's sampling theorem.
Apart from a harmonic algorithm, the realizatior~ of a base al~orithm (6) is
possib le in any other orthogonal base, for example, in the base of the Vilen-
kin-Krestenson step functions. However, preliminary analysis shows that in
such a base there is practically no possibility of reducing computation ex-
penditures in comparison with the algorithm of the correlation of copies.
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It is advisable to use the above-mentioned considerations together with the
obCained results of the synthesis of the algorithms in deve~~pment and analy-
sis of the effectiveness of systems of maximally probable processing of var-
ious radar signals.
Bibliography
1. Khelstrom, K. "Statisticheskaya teoriya obnaruzheniya signalov" [Statis-
tical Theory of Signal Detection], Moscow, 1963.
2, Sakrison, D. "Lektsii ob analogavoy svyazi" [Lectures on Analog Communi-
cation], Mir, Moscow, 1974.
3. Anderson, T. "Vvedeniye v mnogomernyy statisticheskiy analiz" [Introduc-
tion to Multidimensional Statistical Analysis], Fizmatgiz, Moscow, 1963.
4. Frenks, L. "Teoriya signalov" [Theory of Signals], Sov Radio, Moscow,
~
1974.
5. Cooly, J. W.; Lewis, P. A., and Welch, P. D. TRANS IEEE, Vol AU-17, No
2, 1972.
6. Mal'tsev, A. I. "Osnovy lineynoy algebry" [Fundamentals of Linear Alge-
bra], Nauka, Moscow, 1975.
7. ~botnin, A. N., and Strashinin, Ye. E. AVTOMETRIYA [Autometry], No 1,
1975.
8. Gold, B., and Reyder, Ch. "Tsifrovaya obrabotka signalov" [Digital �ro-
cessing of Signals], Sov Radio, 1973.
COPYRIGHT: Radiotekhnika, 1979
10,233
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UDC 621.396.96
~ P,ADAR CONTRAST BETWEEN TWO OBJECTS
~ Kiev IZVESTIYA VYSSHIKH UCHEBNYKH ZAVEDENIY: RADIOELEKTRUNIKA in Russian
~ No 6, 79 signed to press 13 Feb 79 pp 63-67
~Article by A. I. Kozlov~
~Text~ Annotation
Concepts were obtained of dispersion matrices ~MR) of two objects with
one and tne same polarization basis through characteristic values of
these MR and a relationship was found between the ra~~ar contrast and the
type of polarization of the radiating wave.
In a number of cases, the use of polarization effects in radar makes it
po~sible to increase the value of the reflected power considerably by
proper selection of the type of polarization of the radiating electro-
magnetic c:ave. For isolated targets, this gain may reach one-two and even _
three tens of decibels compared. to the traditionally used linear polarized
waves ~l, 2
For example, when scanning remote targets of the earth's surface, of
interest is a comparison between signals reflected from two different
ssctions ~resolution components) for the purpose of selecting a kind of
polarization at which the contrast between them is maximum (by contrast
we mean the ratio between the powers of the indicated signals~.
The given problem can be solved easily if the sought-for ratio is written
using the respective components of dispersion matrices of the two sections,
" hereinafter called objects. However, in this case, it will be necessary
;:o deal with 14 actual parameters: three complex parameters from each `
object and two describing the polarization of the incident wave. In this c
- connection the problem arises of representing the dispersion matrices of
' the two objects on the same polarization bases through several invariants
of these MR. ~n this case, naturally, the purpose is t~ impart to both
MR the simpLest symmetrical form. On Poincare's sphere (Fig. 1), let a
pair of diametrically opposing points (1-1) represent ti~e characteristic
~ polarization basis (PB) of the first object and pair (2-2) of the
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- second. We will designate angle 102, which is invariant to changes of the
bases, by 2oL~Z '
~ ~
~ ~ ,
7 1 \
2a~ ~ ~
_ " zs
,3 r.f
~ Z
I ~
~
%
Fig, 1
, In each of the indicated bases, one of the dispersion matrices is diagonal
A~-i = O1 ~l ~ p2-z = rp' ~l ~
~1 L~ PaJ
_ where ~i and l~Z, pl and p2 characteristic values of dispersion
matrices of the firsf and second objects respectively which, in the general
case, are complex numbers. For definiteness, we will consider that -
~~~~~~~2~~ ~Pi~%~P2~� Indices of and P indicate the polarization
basis~used for consideration.
When changing over from one PB to another, the dispersion matrice, as is well
known E1, 2~ , is subjected to congruent transformation by means of
unitarj~ matrix Q, consisting of faur parameters: b, a,
the general form of which is ~3, 4~ �
~m e`'' cos a- e i~ sin a
e e`~ sin a e`n c~s a~
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If two pairs of diametrically opposite points on Poincare's sphere show
the old and the new basis, then the angle between corresponding diameters
will be equal to half of angle OG ~4~ .
The geometrical interpretation of the remaining parameters ~ ,
and is of nv interest for the problem being considered.
The polarization bases represented by the points on the major circle, the
plane of which is perpendicular to diameter (1-1) in Fig. 1, will be
obtained from PB (1-1) at oC = 11~4 Similarly PB,
represented by points on the major circle, the plane of which is perpendicular
to diameter (2-2), will be obtained from PB (2-2) also at d.= y7~.4. .
The basis which is represented by intersection points of the major circles
considered, we will assign index (0-0) and will call it the zero basis.
The dispersion matrix of the first object on basis ~0-0) will have the
form~
N
Ap_p = QfOAi_~~LlO~ \ i~
and of the second
~
po-o - QaoP~2Q~o� (2)
Indices of Q designate from which to which basis the transfer is being
_ made, for example, Q10 corresponds to transfer from (1-1) to (0-0).
From equalities (1) and (2) follow inverse formulas
~
N
A1-1 = Ql~-o~+G10 ~3~
?
: .
p2-2 = Q20P0-urG20
where the basic property of a unitary matrix
QQ* _ 10 ~
[O1
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is taken into account, while symbols rv and ~ designate
correspondingly operations of transposition and complex conjugation. -
On the other hand, in correspondence with transformations types (1) and
(2) we have:
~
A!-t = QotAo-oQos ~4)
N �
P2-2 = Qozpo-oQoz
a comparison between (3) and (4) indicates that
Qio = Qos, Qso = Q~. (5)
from which it foilows that
ao! = a~o� no! ~ - ~I~o~ ~O~ = n -f- ~~o�
J
' Using relationship (1) and requirement a,o = n/4, we will
write tha form of the dispersion matrix of the first object on basis (0-0)�
1 ~,,e2in:' ~e sr~. ~1ertn~+E~ ~ ~e
~cn~.+~~a
~ Ao-o = - ~6)
2 ~1e~cn..+~~a _ ~e rcn:.+E ~d ~~e~R~. + ~e srn~.
PO-0 has a similar form.
Matrix (6) contains little useful parameters ~~a , ~+a . It is
important to express them by parameters which tie together PB (1-1) and
PB (2-2) , i. e. , angle a(,12 and i2 , -
' ~~z
The transfer from PB (1-1) to PB (2-2) can be made directly or in arrangement
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(1~-1)-.(0-0)->(2-2),
which can be written
~s-s = Qiz~t-~Qiz Qoa ~Qio~i-tQi~ Qoz = QzoQioAt-iQsoQso.
where (5) is used. It follows from here:
N.
Q~ ~ ~i~~r~0 ~ ~~S`01~
which produces~
~lio - ~lao - ~~o -4- g20 = 2al, '
- ~lio - Tlzo ~io - ~so = ~lt~ � . ~
~lio ~Ira ~w ~o = 2~ls n
Relationships (7) represent a system of three equations with four unknowns.
We will write one of its symcnetrical solutions:
ats 'f' ~I~z ~ia _ _ n 1
~!0 - 2 ' 4
~o = ~lia 2z ~ta 4
. ~8~
ais tz ` ~t~ n
~I~o = 2 + 4
_ a~z- ~liz ~~z n -
~o - 2 + 4
We wi11 substitute (a) in the general form (6) for dispersion matrix
on (0-0) basis
n;,.-o = ~F~s - �z) era,. �i -I- �a ~ .
~9)
2 �t ~�t - N~s~ e�-~a`"
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similarly
po-o = t ~S~ - Sx) e ra,. Ss s:
2 S~ Sz (si - s~ eta" . (10)
_ where
N'! = ~,i@~f~�i~~~J
~ _ ~ae ~(~u-FE~,)
S! = pse itn,.-e~.~ .
S~ = plei(n~:-~s~1
Relationships (9) and (10) give the sought for representations of the
dispersion matrices of the two objects in the same PB, expressed by
_ their characteristic values of each MR and the parameters tying together
the characteristic polarizations of each object.
The following representations for Grave's power matrices follow from
relationships (9) and (10); �
1 ~~l'iIZ-~'~"B~Z e ~a~'~~~lIZ-~~,2~2~
G" - 2 era" ~ I~! IZ ~~z ~Z) I~s IZ -f- ~~z ~Z
~ I IP~IZ-I-IPzl2 e`�`'~ (IP~ IZ-IP9I2)
p =
2 e^~`"(IPsIZ-IPZi~ IPsI2-F-IPslt
As may be seen, Grave's matrices depend, besides p and ~ only on
angle OG~Z �
We will now compare the powers of the signals reflected from the objects.
We will write the orthogonal components of the incident elliptically
polarized wave in PB (0-0) ,;.n the form
E cos ~ei~erme
iaeA =
EzR.R = sin ~e-rQ'e~~ . .
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where ~ and ~ determine the form of the polarization ellipse. The _
value of the power reflected from the objects will be determined by
relationships
IIe = EnanGeEnax ~ (11)
~ I
np = EneArJpEts+A
N
where EnaA = (EInaAE2na~.
Relationships (11) make it possible to find the sought-for ratio of the
pawers of the signals reflected from the two objects depending upon the
form of polarization (parameters ,A
1~
II~ a~ 1-~- k~ sin 2~ cos (2~ a,~ -
q IIp v~ 1~- kp sin 2~ cos (2~ - a!~ ' (12)
where coefficients k~ and kp characterize the degree of polarization
anisotropy of the objects
~~ilz-~~z~2 k _ ~Pi~-~Pa~
k'` I~iIZ+I~I2 ' p IPlIZ+IPlIZ '
Q~=I~~IZ+l~IZ. Q~P=IP~IZ+IP$I2.
Sum ~~.~~~-~-~71z~~~ representing a spur of Grave's matrix, is invariant
to bases (1) transfannation. It is equal to the sum of all effective
dispersion cross sections (EPR) with parallel and transverse polarizations
and is called the full EPR of the target (S) (for horizontal and vertical
polarizations Q~=~t.~.-}-Qas-f'2Qra).
External values are assumed by q, as follows fram (12), when sin 2~ _�1.
. In this case:
n 1 Q~ 1 t k~ cos (2~ a~~
4~-912~=f 2 J=
v~ 1 t kp cos (2~ - a!9) '
A further analysis shows that extremums are reached at
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2~ ~ n- arccos kn~, sin 2a�
1~ k~ kp - 2k~kp cos 2a1z )
kp - k~,
at~tg kp + k~ _ tg a1zl -f - 2~t,
I
n=u,f1,... -
It may be seen from (12) that the sought-for value of q represents a
product of two factors, one of which is a ratio of full ~PR of objects
independent of the form of the incident wave polarization, while the
second takes into account the polarization anisotropy of the targets. _
lOR 86
O,B 04 0
6 K =/,0
? -
0,4 0,8 ~
0
2
OS
-6
Q9 Q~
-ro
Fig. 2
1, db
Fig. 2 shows the relationship berween the external values of the second
factor, designated ~
a] R=q Q~� ,
..n
and the value of k~ for various kp , with angle AL~~ assiimed equal
to 7j'~2 . The upper curves in Fig. 2 correspond to the maximum value
of ~t and the lower to the minimum value of R. In other words, when the -
form of the incident wave polarization changes, the ratio of powers of
reflected waves wi11 change from
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Q~~
Rmax to Rmin �
O~ Q~
In conclusion, we will turn our attention to one r~lationship. Let
points (3-3) in Fig. 1 represent a polarization basis for making measure-
ments. The dispersion matrices of objects in this PB will be assumed
- to be S and P. In this case, for It~ and ~p the following ~1, 2]
is true:
k~= V 1-4~dets~ kp=, /-1-4~detP~.
a~ ~ Q~v
If an~les pG13 and pC~3 are introduced as was done in Fig. 1, then
- o~,, cos 2a, ' kp vi~ cos 2a~' ~
where Q'~~ and ~22 are the .EPR of two orthogonal polarizations.
BIBLIOGRAPHY
1. Kanareykin, D. B.; Pavlov, N. F.; Pote?.hin, V. A. "Polarization of
Radar Signals." Moscow, SOVETSKOYE RADIO, 1966
2. Kanareykin, D. B.; Potekhin, V. A.; Shishkin, I. F. "Marine Polarimetry."
Leningrad, "Sudostroyeniye," 1968.
3. Gorshkov, M. M. "Ellipsometry." Moscow, SOVETSK~)YE RADIO, 1974.
4. Kozlov, A. I.; Demidov, Yu. M. "Certain ProperYies of Covariatianal
Dispersion Matrices." RADIOTEI~INIKA I EI~EKTRON~KA, 1976, 21., No 11,
p 2415.
COPYRIGHT: "IZVESTIYA WZOV SSSR-RADIOELERTRONIKA", 1979
[269-2291)
- 2291
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UDC 621.396.9b
THEORY OF NOISE-LIKE SPACE-TIME SIGNALS
Kiev IZVESTIYA VYSSHIKH UCHEBPZYKH ZAVEDENIY: RADIOELFKTRONIKA in Russian
No 7, 79 signed to press 28 Apr 78, after revision, 3 Nov 78 pp 3-10
~Article by A. I. Pogorelov~
~Text~ Annotation
A definition of noise-like space-time signals is given and their classifi-
cation is described. Problems of forming and processing phase-manipulated
space-time signals were considered. An analysis is given of the output
effect of an optimal system for processing such signals.
Space-time presentation of signals was found to be very fruitful in
~ solving a great number of problems in radar, navigation, control and
communications. At present, it has achieved a level where problems in
the practical implementation of the aptimal methods for transmitting
data and measuring motion parameters can be solved successfully. However,
unlike systems with "time" signals, the optimization and practical
implementation of which are being developed in the direction of seeking
the optimal structure of the signals, as well as finding optimal algorithms
for their processing, in systems with space-time signals the solution
of these problems is limited at present, as a rule, only to searching
for tYie optimal procedure for processing space-time signals (PVS). This
narrows consid~rably the area of using the space-time approach and makes
it impossible to unveil and implement its possibilities fully because
_ the problems of selecting the shape of the signal that determines the
system structure, as well as its basic indicators, are some of the most
important problems being solved in ita design.
In synthesizing optimal time signals, attention was given to pseudorandom
- or noise~iike signals (ShPS). A number of properties that were responsible
for their wide dissemination and that predet~:rmined great posaibilities of
the systems when they are used ~1~ , provide a basis to consider that
many problecna of ineasuring motion parameters and transmitting data may
be solved by using noise-like space-time signals (ShPVS).
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Below problems of forming and processing ShPVS are considered for viewing
a given comparatively wide sector of space.
The PVS may be represented in the form of a vertical or horizontal vector
of elect~ ic field intensity with an angular density s(t; where
6 angular coord.inates of the PVS arrival. direction. In the
simplest case we will consider the PVS with a single-dimensional angular
density ~ i 9 . In this case, like the pse:udorandom coding
of signal s(t; with respect to time, it is possible, by taking into -
~ account several limitations, to implement pseud.orandom coding s(t; ~ )
and along space coordinate e , where e = sin'~`
angular coordinate. Such PVS s(t; which has a noise-like structure
along time, as well as space coordinates, we will call a noise-like space-
time signal.
Like time noise-like signals, the ShPVS may be classified into signals
with analog and discrete modulations, which may be formed as a result of
their space-time modulation at intervals ~tj !F T and ~e~` eM �
In case analog modulation method~ are used, the following expression can
be written for the ShPVS:
s (f; e) = S (f; exp {I (~ot -f- ~o)}~
,
where S(t; e)-- complex envelope of signal
S(t; 6) _ ~ S(t; 6) ~ exp {j~ (t; 9)},
determined by the kind of modulation used, while ~o initial phase.
Any methods of time and space discrete coding can be used in principle
for signals with discrete modulation. As an example, we will describe a
discrete-coded signal using symmetrical cyclical coding which leads
to some simplification of calculations, as well as simplifications of
arrangements for practical implementation of systems with the ShPVS. In
this case, modulation with respect to time t and space 6r is done according
to one and the same pseudorandom law, for example, according to the
law of some pseudorandom sequence (PSP) which, in the case of using a
linear antenna array, leads to the foll~wing expression for a discrete-
Coded signal~
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,o ao en n
s~t' 9~ St ~P {I I~~o wd t-I- Wl1} X
t=-~ p=-eo k=-M t~--n
x d~II f t-(l ~ f- i2T I ff ~e) n~9 - k0 e 1~~ 1. ~1)
t. ~ ~
where N-- number of PSP elements, while d.e and ~t respectively
' values of the space and time elements of the PSP; S~, c~c, ~t
respectively values of the amplitude, frequency and phase in the 1-th
position; dl =+1, -1, f(~)-- determines tY~e forni of space PSP,
(7 ( X cutting ~unction detenuined in the following manner:
1? Ix) < 2 ~
II (x) _
. 0~ Izl> 2 .
2m 1 = �e" = 2n 1 = ~ = N, (2)
� At = ~9 eY . ~ (g~
In this case, the discrete-coded signals may be classified into amplitude-
manipulated, phase-manipulated and frequency-manipulated, as well as
signals with combined manipulation. The structure of phase-manipulated .
. ShPVS when using for the space-time coding a seven-element Barker code
(-I--~+--1-) is shown in Fig. 1.
Fottning a noise-like PVS structure with time coordinates is not difficult
1. At the same time, in space coding, for example, it is difficult
to form the required rectangular shape of the space element of the pseudo-
random code fk (e which leads to a peculiar space structure of the signal
that differs from that ideal ShPVS structure shown in Fig. 1. These ~
difficulties are caused basically by the limitation of the space frequency
spectrum, due to the limited dimensione of the aetual aperture of antenna
systems.
I
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-4- -F' - - _
- - -t- -
- - - -t-
B
- - -
-~---~-F-~'-
- ~ - + - _
- -i- -
t
.
Fig. 1.
We will now consider the special features of noise-like space-time coding
in which we will assume a given value 2x,~=2X,~/~,~ of the
space frequency spectrum, where 2X maximum linear aperture of the
antenna system, while - working wavelength.
To form the ShPVS, we will utilize the we11-known methods of forming
complicated antenna radiation patterns (DN) for which the product of the
width cf space frequency spectrum 2x~�r by the "duration" of space
interval 29K scanned is considerably greater than unity [2, 3] .
" The formation of such DN, used basically in processing PVS, is based
on the space-time modulation of the amplitude-phase distribution in the
aperature of the antenna array. In this case, the formed PV5 s(t; B )
, may be represented thus:
s (t; 9) = s (t) F (9; (4)
where F( 8; t) DN, determined by the amplitude-phase distribution _
I( x;t) in the well-known way
xy
F(9; S 1(x; t) exp j2nE~} dx. (5) -
-xN ~
As follows from (4), the space characteristics of the signal are fully
determined by form of DN F( 9; t) . Substituting DNin the fona
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F(6; ~ F(E~; t) ~ exp {icp (9; t)} (6)
and formulating the condition for a unifox~n scan of the given space sector
QM ; e~�~ ) in the form
J.F (8; t) Fa = const, I 6 gx , ~7~
we, taking into account (5) -(7) and for a condition 28y`XY 1 ,
srri.ve at the following expression for DN:
M
F(6; t) Fo exp { jcp (k06; t)} sin c 2n7(� (9 - k~6), (8)
k~-m
where
sinc z ~ sizx ~ 09 = 2K� ; 2m I= 29�2x,,.
Deteztnining ~Q(~r06; 1) as cyclic, 2T-periodic functions
differing from each other only by a shift in the integral number of time
intervals d t, the number of which in interval 2T we select equal to
the number of DN (8) discretization points in interval 2 8M
c~ (k06; t) _ ~o (t - k~9 ~ - i2T) , i = - oo, , oo, .
- we obtain from (8)
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ELt~ ~ i ~ ' I CRL ENG ' I NG

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~
r T
F(~3; i) = Fa exp ~l �o I t- 6 e~ - i2T
\
o_�
' oo m \
_ Fo eXP j 1~01 f- k~~ e~ - i2T j~ sinc 2n~ f~ - k~9). (9)
~ ~ 1
- r=-~ a~-r,~
The structural arrangement of the system that implements the algorithm
for forming the ShPVS (4) may be represented in the form of a linear
antenna array shown in Fig. 2, the radiation pattern f~rming ttie arrange-
ment (DOS) of type
m
F (6) _ ~ sinc 2nYa (0 - k~6),
k a-1fl
Modulators (M) are installed at the input at each of its channels, c~ntrolled
by signals of a modulating function oscillator (Ghi~') and performing in the
general case frequency, as well as phase modulation uf signals s(t)
- received through distrihution device ~RU) from the high frequency
osciilator (GVCh).
QOC ~ 1 ~
, M 2
_ M ~~v~
M
~3) Py ~By
(4)
F'ig. 2 ,
1. DOS . 3. RU ~
- 2. GMF 4. GVCh
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When utilizing discrete-coded ShPVS, functions ~(kAO; t)
will be represented in the form of functions quantized by time and
levels. Of great pracCical interest are functions ~(k09; t).
that have only two quantizing levels, for example, 0 and the
quantizing with respect to time of which is done according to a law for
- some PSP with 2n + 1 elements and a duration of (2n + 1)`C~ = 2T.
~n this case
~ ~n~ ~ ~ 0
exp {i~o (t - k06 Y- i2T11= ~ fi/ din r t- l k T- i2T ~
` ~ T~ J
!m_ao [~-n
and the formed phase-manipulated ShPVS has the foztn
0o rn n
S ~t' e~ - ~ ~ ~ s~el~~d~~ Lt - (1 +ti ) tio- i2T1 sin c2nX� (6--k0A).
o ~
t=._,s (10)
The structure of signal (10) is very unusual, Thus, in any of fixed
directions ~k- k09, coinciding with the DN discretization point,
there is formed a signal
co n
S(t> e) S e~~'`d II( t'~~ k~ ~ro - i2T 1
a c ~ z
t~-~ t~-n �
- differing from signals formed in other directions of dis~retization
"�e,: f I v.,x I~V dv..:~
~=~-, x=x-.
2. Angular Field Energy Spectrum and Distribution Parameters -
We will nuw determine UES taking into account the special features
characteristic of near radar problems. As is well known, the field
above the interface surface within direct si~ht limits is the sum of
two components. The first component corresponds to the wave propagation
_ reradiated by each element of fhe multiplexity that forms the target
in free space; the second component is formed as a result of their
scattering by the interface surface. Considering the case of a scalar
field, it is possible to represent the latter in the vicinity of the
observation point located at the start of the coordinates (Fig. 1)
in the following form:
the field created directly by the target
Ue~ = a,; (r~, ~r) UT et~~~+k~ir
l�
S 2 y-~.~ ,
~
the field reradiated by the interface surface
a~ ~ra, 'Kc~~ Uc ~r~~ ei(kr'+k(ri)r~~
(86) l1R, = S 2 Yro,~ -
b
-~l ~
wheri k(r~~s) _-k( c,s/r~~s) wave vectors of elementary waves;
UT (r~) field irradiating the target, containing the random, as well
as the determined components; U(r field created on the interface
c s
surface by the target; a~( c, ~s~ s~ ~cs complex scattering
coefficients of target volume elements and of the target surface depending _
upon the directions af radiation and reradiation.
� 209
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_ Integration in the first component is made over the area of the space
containing the reradiating elements of the target and in the second
_ component over the entire surface of the interface. We will note
that the determined component is contained only in (8a) and it is
equal to
~8~ Ux AeT9 1 Us An ~1'eo) aeo ~74co~ kr) ~ T~1, eT = Tdet
2Yn r~o
~ where UTdet ~r~) determined component of the field UT( radiating
- the target. Assuming that adjacent reradiators of the target and the
interface surface are not correlated, i.e.,
=o ~or T~~-`Tsr
- we will obtain, averaging over the set of numbers, the following
expression for the correlation function of the field component ;st the
reception point:
s
_ < U,~~ ~P) >-i-
U~~a~ (P) >-L
R A.n~ R Aes�
Then changing over to polar coordinates and making the Fourier�trans-
formations, we will determine the energy spectruni:
(9) s ~~P~ x) =S~ ~q~~ x) +s. x) ~
_ ?f
(~a, UQ \TI h~ a Ci~$ ~ r 1' Q~ \j.e1 ~'T~ ~
n
_ .
XSs~r'a~ (11'+Oo \~eo~ xr~~s \Teo~ S~3t-3tp~ J r
(96) s� X) ~ 1 p: (T.) S. (t.~ T~~) X.
n
Xctg x1' 1+tgz x(cos ~-sin 2,
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where cfa = 1/2 a~oa~a effective scattering surface (EPR) of the
determined reradiator; d= 1/2 ~ a a~ unit ~olume of the EPR
of the target; Ps ~ s~ � 1~2 ~ as 8s~ utiit coefficient of diffuse
scattering by the interface surface; ST (r 1/2 < U TUT average -
density of the total flow of radiation to ~he target; ST ~rco~
average density of the random component of target radiati4n; S(r ,r -
average density of the total flow of radiation of the interface surface
from the target.
For the case most frequently encountered in practice of target dimensions
being small compared to the distance to the scattering region of the
surface
_ ~10~ s~ \Ty T~e~ ~~f7tT = [ J ST `Te~Q~ \Te~ xsi xc~~dV'~'
io
9~
+S: (ra.) ao (~zr ~a.) J .
It may be seen from (9) and (10) that in the general case the angular .
energy spectrum at the reception point and, therefore, the parameters
of the angular coordinates distribution law (1) related to it depend
on the spatial structure of the rad~ating field (function ST(~c), as
well as the averaged indicatrix of the target reradiation.
We will now evaluate parameters (3) -(7) at situations characteristic
for the SBRL. Since the vertical dimension of the target, as a rule,
is considerably smaller than its height above the surface, on the basis
of papers ~ 6, 7~ , the radiation field of the target at diffuse
scattering on the transmitter-target route may be considered uniform
and it may be assumed that ST = const. Then parameter Po, which is the
ratio of the determined field component power to the average flow of
the random power component, wi11 be
~a AesUB Aes Poe
~(11) S(~'x)~dx 1+(1+Pa) (Ra~-}'Ra-f-Re~Ra) '
JJ
0
ao
~42) Po~= .
f �'(T`)`~'
qe
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here
�~j13) Raa P: ~r.) ctB' %y 1-t8?X (cos W-sin -
4i~r.a
Q
is, according to the terminology of gapers ~2, 6~ the di�fuse
scattering coefficient, although it is more correct to call it the
energy coefficient of diffuse scattering.
Parameter RaT, determined as
. y
S~ ~r~)
d 5:~~'a~-~s~Te~ ~
is obviously a similar characteristic for the transmission-target route.
Thus, the value of Po a~ the reception point differs from the similar
characteristic of the target (12) when there is not interface surface.
In particular, for a characteristic, according to ~6] , for a wide
region of angles of slide Rd = RdT = 0.2, even for a fully determined
reradiation of the target (i.e., Po~ = oo ) the value P = 2.3,
i.e., the determined component of the power is only slight~y higher
than the random component. For RaT = o and P which, in
particular, corresponds to measuring the coor~inates of the radiation
source, we obtain P,=R~'' and for the previous Ra = 0.2 now Po = 5,
i.e., it is u~ore than double the value in the radar case.
Suhstituting (9) and (10) into (5), we will determine the elevation -
coordinate of the radio brightness center of the reradiating region:
; (14) x�,a xa ~1+R~)-t'XoPoeR,~-1-x,'Rati+R~) ~i-FPo~) .
i+(1+Po~) (Rd+R~-~R
;f
f xS~,. x) ~dx
a
~i~~ ~ -
7La.~~ ,
. j~ s.., x~ ~dx
n
where ;~o coordinate of the determined reradiator. In the parti-
- cular case of Po~ = o, the coordinate distribution 1aw ~�1) is symmetrical
212
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with respect to x and, therefore, the latter is an average coordinate -
of the i:nstantaneous radar center of the target, i.e. ,
Its value in free space (i.e., Rd = Rd~ - o) ia equal to .
=x~ ,
and when an interface surface is present
(i5) = 1+Rd~~~'+'Rdx,').
~
Since xs is always negative, it follows that the MRTs is shifted
down due to the effect of the diffuse component of the average coordinate
as compared to the case of free space by value, from (15~
(ig) nx=-=(x,'-x.').
1+Rd
For Poy ~ o, the value of > must be determined by numerical inte-
gration of ~1); this case will be analyzed later. Substituting (9)
and (10) into (3) we will obtain, taking into account equality
11 ~x-x') S(~p, Y.) d~dx~ XZS 7G) d~dx- ,
0 0
-~x'~' Jf S c~, x> a~ax
0
the following expression for parameter ~ Z ,
x
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(i?) E,,xZ=
(�K=+~x~ ~s+~xo~= PooRa~+�.x'Ra~1+Po~)+~x.')ZRa~1+�oa), ~1+Rd:)
~ 1+Ra:
_
_~x~~=~ 1-i-(1+Po~) (Rd+Ra:+RaRar) .
~x-x~ s~,. X ) d~dX
o '
(17a) .
J J s~,. x) d~dx .
~ -
For the azimuthal coordinates of the relationship, equivalents (14) -(17)
are obtained by the simple replacement in the latter of variable ~L by
Differences in results originate when special features of the
angular energy spectrum are taken into account. In particular, for
isotropic roughness in the interface surface, the energy spectrum with a
point target is syuQaetrical with respect to plane Cp .
Because of this , i,e., no shift occurs of the averaged
coordinate of the MRTs. At Po~ = o, we obtain
- (18) E4~=~l~aZ~'I~.o=Re) ~1+Ra)''.
3. Quantitative Evaluation of Values
Thus, the calculation of statistical characteristics of angular coordi-
nates with previous assumptions is reduced to finding parameters
~1 x, c~#, x*~ for the target proper and the interface surface.
To est~ablish the basic laws, we will evaluate the relationship between
these parameters and the relationship berween quantitative distribution .
characteristic (1) and the route geometry and the value of Rd. In the
calculations, we will assume a point target at height h, the same as
that of the receiver, and lying in the coordinate plane (i.e.,~ = o).
The distribution law for the slopes of the irregularities we equally
probably with maximum valuea of `rmax = 0.1. Calculation results of
dependences of paran~etera ~1,,q~ ~t~, ~ Ixs+ on the ratio of the
height of the corresponding points to the distance D between them is
shown in Fig. 2a. It may be seen from it that parameter xs (curve 1) -
is always greater than the angle if mirror reflection from the average
plane (broken straight line), which indicates the prevailing role of the
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half of the route nearest to the receiver, Parameter ~isx (curve 2)
remains almoat unchanged in a wide range of h/D ratios, near their ~
asymptotic value of 0.065. Parameter ~NJ~ (curve 3) changes almost2
linearly with an increase in h/D from its initial value near 0.81/ ~ ~
We will note that for a normal distribution law of irregularity slopesX~
the nature of the parameter changea remaina the same as shown in Fig. 2a,
and by selecting the effective value of the slopes properly, it is _
possible to obtain a good quantit.stive agreement. The relationship
between ~us4 and h/D makea it possible to make more preciae ~he concept -
of "point target." If the following inequality is assumed as a criterion
NY.< ~,.~0,8i~r:.o~
then we will obtain for the angular size of the target, with an equally
probably distribution cf unit EPR along angle
e.=zvs�~. and the value of Rd for cases of a fully
random target (solid lines) and a radiation source (broken lines). Similar
_ relationships for azimuthal coordinates are shown in Fig. 3b. Curves
: in Figs. 3a, b were plotted for h/D = 0.05, ~mBX = 0.1.
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~ IoXI,