ELECTRICAL COMMUNICATIONS

• Document Type: CREST
• Collection: General CIA Records
• Document Number (FOIA) /ESDN (CREST): CIA-RDP81-01043R002000220003-8
• Release Decision: RIPPUB
• Original Classification: K
• Document Page Count: 132
• Document Creation Date: December 27, 2016
• Document Release Date: June 3, 2013
• Sequence Number: 3
• Publication Date: January 1, 1957
• Content Type: REPORT

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STAT
AIR TECHNICAL INTELLIGENCE
TRANSLATION
STAT
id
STAT
STAT
'ELECTRICAL COMMUNICATIONS
(ELEKTROSVYAZ')
NO. 3, 1957
PP? 1-80
SVYAZ'IZDAT - MOSCOW
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0,
Table of Contents
Page
The Noiseproof Feature of Receivers with End Recovery Time, 1
by A.M.Vasiltyev ..........................................
Generalized Analysis of Amplifier Stages, by A.A.Rizkin ........... 8
Determination of the Basic Parameters of Multichannel Radio
Relay System Apparatus, by S.V.Borodich ... ................ 17
An Investigation of the Self-Oscillating System in an Oscillator
with a Semiconductor Junction-Type Triode, by S.M.
Gerasimov ................................................. 33
Frequency Band of Radiotelegraph Signal Transmissions with
Amplitude and Frequency Keying, by M.S.Gurevich ........... 50
Contactless Switches, using a Transformer Ferroresonant
Circuit, by Ya.G.Koblentz, D.A.Yakovenko .................. 63
Methods of Phantom Telephone ConIInunication, by N.V.Reshetnikov .... 79
A Polytonic System of Number-Transmission in Long-Distance 89
Channels, by V.N.Zachesov .................................
Calculation of the Output Capacity of Telegraph Service
Equipment, by P.V.Prakhov ................................. 97
From Foreign Journals - A Radio-Repay System for Television 115
Transmission ...............?.............................?
Foreign Patents ................................................... 122
0
New Books ......................................................... 127
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,-output or when receiving strong signals which overload the receiver, the end recov- }
ery time must be taken into account.
It is of interest to define the probability p(t)dt of receiving a signal in the-
-time interval from t to t + dt, considering as given the probability n(t)dt of the
-arrival of the signal at the receiver input during the time interval from t to t + dt.
.0
Below, the arrival of a signal at the receiver input, during the time interval adja-
cent to t + dt, will be considered an independent event. This corresponds to an ap-
proximation when "the infinitely small" interval dt is denoted as the time interval
of the pulse duration series, and n(t)dt as the probability of the appearance of a
signal for an interval of time approaching the point t.
Equation for the Degree of Probability of Signal Reception
We will now consider the equation for the degree of probability p(t) of signal
reception in the receiver with end recovery time T. The probability of signal re-
ception in the interval t - t + dt is equal to the probability of signal reception
at the receiver input in the interval n(t)dt, multiplied by the probability that the.
0
receiver is in recovery condition. It is obvious that the receiver will be closed
if the signal is received at any of the instants of time 9, lying in the interval
t - T < E < t. The probability that the signal is received for the time t - T to t
t
is equal to f p(E)dg. Therefore the probability that the receiver is in recovery
t t 0 condition at the instant t is equal to 1 - f p(F,)dp-. It is obvious that the
t -T t
probability p(t)dt may be represented as the product n(t)dt [1 - f p(g)dE].
t -T
Equating both expressions and reducing by dt, we get the desired equation
1
P (t)= (t) 1 - f p (c) do
r
(1)
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If the signal received is transmitted periodically at-fixed period T and inter-,
f then the degree of probability of signal appearance at the receiver output
n(t) has the constant value it for all instants of time when the signal is not being
transmitted and is rejected during the instants of time t = kT when the signal is
being transmitted. These rejects correspond to the fact that, at these instants,
the probability of signal appearance at the receiver input increases- sharply.
If S(t) denotes the delta function, determined by the conditions
+ 1
0 1 -/= d1, ~11
6(t)dl M t0
-(1)-- q' U),
is
(2)
where q is the probability of the appearance of a transmitted signal at the receiver
input in the presence of interference. Thus the determination of the degree of
probability of signal reception at the instant of time t, if a periodic pulse signal
is transmitted and the receiver is affected by a fixed interference, reduces to the
derivation of eq.(1) when n(t) is given by eq.(2). If reception begins long before
the instant of time under consideration, the solution of eq.(l) is hardly distin-
guishable from the fixed derivation, i.e., from'the periodic solution of eq.(l),
with n(t) as given by - cc < t < + w. In the following, only the periodic solution
of the equation will be considered.
-Solution of the Equation for Low-Intensity Interference
The solution of the equation under ordinary circumstances is far from simple.
In this connection, the case below which the intensity of interference is low will
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then, for n (t) in the case under consideration, the following analysis may be writ-
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be considered. Interference may be considered low if n't ? 1, i.e., if the probabil-{
ity of a false signal arriving at the time T is much less than unity. It is natural
to limit the calculation to the case where the recovery time of the receiver T is
less than the period of signal transmission T. since otherwise, even without inter-
ference, not all of the transmitted signals could be picked up by the receiver.
We will represent the degree of probability n(t) as n(t) = it + no, where no =
= E q g(t - kT), and the degree of probability p(t) represents the analysis p(t) _
k
= po + pl + P2 + ..., where pn is a member of the series n conforming toaT. If
these terms in eq.(l) are substituted and the members of one finite series equated,
then a systems of equations is obtained which permits the determination of all pn:
Po - -0
1 - J Po (E) dE
1 1-7
P1 . 1 - J Po (E) d-' -- -o P i (E} dE
- r= r--
r
P2=-z $i()d_o SP2)do
t t
PR = -zr f P?-1(E) dE -- rro f p,,
(E) d;.
po(g)dg = 0,
kT ~- T
The first equation of the system (3) corresponds to zero approximation, i.e.,
to the case where it equals zero. Only at t ~ kT no = 0 does it follow that
at t / kT. At the points t = kT n o = q 6(t - kT)t = kT,, and the interval
(3)
Po =
since it takes into its scope where po(e) = 0 (g / kt). From
this, it is obvious that po = no.
?
A study of the second equation of the system (3) indicates clearly that it cor-
responds, in first approximation, to nT. In following this through as po = Ito, this
4
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1 1
takes the -form of P i ( ) 1 o d` Tro pi (c) d'c,' at t ~ I:T, Ro = 0 and, conse-
quently,
I
Pi (t)=7C 1 - S710 (E)d`
l'
(4)
where, in conformity- with the symbols introduced previously (Bibl.l), we use
direct (I) = i
For the points t = kT, we may now write
kT kT
p i (kT) = I- c1o -- q% (1 = kT),_kT J Pt (E) d'
kT kT-T
or, neglecting the first term on the right-hand side versus infinity, as the value
of the second term and considering eq.(h),
F kT
Pt (k T) q J r.. - rq direct
kT--
'r
Let 'c > 2
; then,
0 jtj >
(_kr_-)d
'c
(1- kT)1-kT?
dips(kT)=-q[,::--q(2---T)]=---q,IT -(2q- 1).].
r inall. -,,.,e obtain for pl(t )
P1 rq direct
k
direct
l -i
IT - (2q kT).
(5)
The following appror._mations may be derived in an analogous manner. It can be
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demonstrated that, if nT < 1, then the series p(t) = po + pl + p2 + ... converges
uniformly; and, consequently, represents the solution of eq.(1). 1:'e are limiting the
calculation to the solution in first approximation which, according to the discussed
? principles, has the form
p0+P1=Jq(I -?[T--(2q-I)(f-kT)+
k
(6)
The Relationship of the Number of Correct Signals Received to the Number of Spurious
Signals
The electronic receiver admits all signals arriving at its input. Therefore,
0
if the degree of probability of the arrival of signals at its input is given in
eq.(2), then the relation of the number of correctly received signals to the number
of spurious signal is equal to --T.
in the receiver with end recovery time, the degree of probability of signal re-
ception, in first approximation, is described by eq.(6). The probability of recep-
tion of a transmitted signal, according to this formula, equals
q{1 -;.[T-(2q- 1)z]}.
The mean number of spurious signals picked up by the receiver `or a period T.
is equal to
kT-I1
S (Po-1 P1)dE=..;(I q)-''--(T-~)=TrT[I-qfl.
kT-T+U
11hence the desired relationship in the receiver with end recovery time will
t-kT- 2
+ T - zrq direct
k
is
equal
q (I -ft IT -(2q- I) r1l.
aT [I - tgT I
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and is distinguished from the corresponding relationship in the electronic receiver
by the comon multiple
d= (t -n[T--(2g-1)t1)
It-~T I
For example, if q = 0.95, T = Q, T. and nT = 0.1, then d = l,.. The mean number
of correct signals in the electronic receiver in the case under discussion is 950,
whereas in the receiver with end recovery time, it is q{1 - n[T - (2q - 1)T]}, i.e.,
93`x. It is clear from this that, in the given case, the end recovery time results
in practically no change in the number of correctly received signals, but leads to a
quadruple reduction in the number of spurious signals.
1. Vudvord,F.M. - Information and Probability Theory in their Application to Radar.
Publ. Sovetskoye Radio (1955)
Article received by the Editors 29 May 1956.
6 ~
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GENERALIZED ANALYSIS OF AMPLIFIER -STAGES
by
A.A.Rizkin
Generalized systems and generalized matrixes for various
types of tube and semiconductor stages are presented. The ap-
plication of the theory to the analysis of equivalent' systems of
semiconductor stages at high frequencies is demonstrated.
Generalized Systems and Generalized Matrixes
The analysis of amplifier stages (tubes or semiconductors) in a linear hookup
is ordi'arily made by constructing an equivalent corresponding system in the form of
a linearly operating quadripole, containing dependent sources of current or voltage.
For the analysis of complex multistage systems, the matrix method is especially dell;
suited.
In addition to this, it is desirable to obtain a separate stage which will be
adaptable to stages of various types. For example, in the case of stages in elec-
tron tubes, it is desirable to have a generalized matrix, uniformly suited to stages;
with a common cathode, grid, or anode; and in the case of stages in semiconductor
triodes, to stages with a common emitter, base, or collector.
This can be easily attained if the generalized equivalent systems of amplifier
stages with double dependent sources are introduced into the types-presented_by the
author (Bibl.l). One of the possible variants of the generalized equivalent circuit
of the tube stage is given in Fig.l, and the semiconductor stage in Fig.2. The
transition from the generalized stage systems to the concrete types has been carried
out in conformity with the data of Tables 1 and 2.
The generalized equivalent systems permit the corresponding generalized matrixes
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to be obtained. Thus, the system of equations for the voltage nodes of the circuit
in Fig.l directly determines the matrix [y] of this system, whereas the system of
Fig.2
The generalized matrixes of-tube or semicon-
n7.1, , ductor stages obtained in this manner are given in
S
Tables 3 and 4. To obtain the matrixes of the con-
crete'stages it is sufficient to substitute, for
the corresponding generalized matrix, the values
conforming to Tables 1 ?and, 2.
According to the known matrixes of the individual stages, the matrixes of the
complex systems are determined. These permit obtaining all values which character-
fSU,
equations for the current circuit of the diagram in
Fig.2 is the generalized matrix [z] of a semicon-
ductor stage. The generalized matrixes of other
types are obtained from these matrixes by the known
conversion formulas (Bibl.2).
Fig.l
ize the operation of the system.
If a source with an emf of Eo and,an internal resistance of Zo operates at the
input of the system, while the output of
I, Iz the system is closed at the resistance Zn
Fig-3
UEf `" (Fig-3), then the amplification of the sys-
tem through the voltage Ki, the amplifica-
tion of the system through the current KJ,
the input resistance of the system Zin and its output resistance Zout can easily be
found according to the formula presented in Table 5 (Bibl.2).
In compiling the Tables for positively directed currents and voltages, the di-
rections shown in the diagrams were assumed, whereas the matrix specifications cor-
respond to the following matrix equations:
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1111
2J - [yl Eu2] ' [U2] = [ZJ ! j' J ;
EU'' = t here m - 0. i.e.. excluding the common collector system.
Using the data given in Table 2 for the common base system and the formulas
BIBLIOGRAPHY
1. Rizkin,A.A. - Generalized Theory of Tube and Semiconductor Amplifiers. Elektros-
vyazt No.1 (1956)
16a
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2. Zelyakh,E.V. - Fundamentals of the General 't'heory of Linear Electrical Systems.
Izd. AN SSSR (1951)
3. Vasseur,J.P. - Annales de Radioelectricite, April 1956
4. Migulin,I.N. - Equivalent Systems and Parameters of Plane Semiconductor Triodes.
Elektrosvyazt No.9 (1956)
Article received by the Editors, 8 October 1956.
16b
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DETERMINATION OF THE BASIC PARAMETERS OF MULTICHANNEL RADIO RELAY
SYSTEM APPARATUS
by
S.V.Borodich
The method for determining the optimum values for the basic
parameters of multichannel radio relay systems with frequency di-
vision multiplex and frequency modulation is discussed. The op=
timum quantitative correlations are established between the indi-
vidual noise components in the telephone channel.
.. _Introduction
In designing multichannel radio-relay systems, the designer must primarily de-
termine its basic parameters, based on the fact that the equipment must supply the
? required quality of the coupling, and that its practical execution has to be con-
nected with minimum technical difficulties. In other words, the designer must do-
termine the optimum values of the basic parameters of the equipment, including:
power of the transmitter; antenna amplification factor; receiver noise factor; fre-
quency swing of the transmitter within the channel; maximum amplification of the
tandem office; parameters, determining the nonlinearity of the line route (linearity
of the modulator, demodulator, and the phase characteristic of the channel). Several
of these parameters may be selected on the basis of existing technical possibilities;
however, in order to make a proper choice, the designer must know how any change in'
one parameter will affect the others.
The present article is an attempt to establish the method of determining opti-
mum values of the basic parameters of the equipment. In summary, this article. has
0 been included as a supplement,to the new recommendation to the MKKR [International
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-Consultant Committee for Radio (Bibl.l)], to determine the allarrable noise intensity
at the end of a hypothetical standard circuit for the radio-relay cocmmnications
systems. The recommendation was accepted by the Eighth Plenary Meeting of the MM
in September 1956.
Initial Data for the Solution of the Problem
The basic quality factor of communication in a telephone channel is its noise
level. All values higher than the parameter of the equipment immediately influence
the quantity of noise in the channel, whereas the remaining characteristics of the
telephone channel, such as stability of overall circuit attenuation, frequency and
amplitude characteristics, transit time of the signal, proofing against audible
crosstalk, channel stability, and so on, do not depend on these parameters. The
noise volume in the channel depends also on the length of line, its construction,
and the conditions of radio-wave propagation. Consequently, the allowable noise
volume serves as a guide for the designers at the end of a hypothetical standard
circuit, which possesses a determined length and structure.
New recommendations by the MKKR, adopted at the Eighth Plenary Meeting, define
two hypothetical standard circuits for radio-relay systems and give the permissible
noise level at the ends of these circuits (Bibl.2).
The first hypothetical standard circuit is designated for radio-relay systems
with a capacity of 12 - 60 telephone channels. It has a length of 2500 km and con-
tains three pairs of individual, six pairs of group, and six pairs of supergroup con-
verters, as well as six pairs of radio modulators and demodulators, respectively.
This circuit is divided into six sections of equal length, contained between the
stations with signal demodulation.
The second hypothetical standard circuit, designated for radio-relay systems,
with a capacity of more than 60 channels, has the same length and number of indi-
vidual and group converters, but contains nine pairs of supergroup converters and
18
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-;nine pairs of radio modulators and demodulators. It is divided into nine sections
01
-Iof equal length, contained between the stations with.signal demodulation_and-modU1&
_'tion.
The common psophometric noise power at the relative zero level in any telephone
-:channel, at the and of both the first and the second hypothetical standard circuits,
.is determined by the following time values (Bibl.l):
a) Average power per hour of greatest usage; in the absence of fading, this
may not exceed 5000 ??w;
b) Average power at any hour, not to exceed 7500 ??w.
These values make no allowance for the noise of the multiplexing equipment
?
46
whose poker, in conformity with the NKKF recommendation, does not exceed 2500 ??x.
It should be mentioned that this recommendation, strictly speaking, is not ado-!'
quate. At the time of strong signal-fading in the course of short time intervals,
the noise volume may grow so much that the connection is broken despite the fact
that the average noise volume per hour does not rise above the permissible values.
The recommendation, therefore, indicates that it is essential to determine.,the man-
ner in which the allowable noise level should be assigned in the course of short
time intervals; however, at present, it is impossible to give a precise answer to
this question.
The above-mentioned recommendations by the MKKR which define the hypothetical
standard circuit and the allowable noise volume, may be used as initial data for de-i
termining the equipment parameters.
OptimumRatios of Noise Components in Telephone Channels
- It was shown in another paper (Bibl.3), that the psophometric noise magnitude
in the telephone channel of radio-relay systems (without considering the multiplex-
ing equipment noise), at the point of zero relative level, equals
P Pj. +Pn.2+Pnns=A fax+Cx2 Nlsw,
(1)
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(Bibl.3) by the following expressions:
GA is the antenna amplification factor (relative to the unipole)~;
Ro is the length of one stretch of the line;
a is the transmission line fading, in nepers per unit length;
where n is the coefficient of receiver noise;
ptr is the power of the transmitter (in ??x);
AFk is the bandwidth of the channel (in cycles);
kn is the psophometric coefficient;
Fk is the mean frequency of the channel in the linear spectrum;
X is the wavelength;
is the entire length of the feeders in one section.
B ~ _ ,Fkc1F tUt' e4b j 2 (ak) [4e_2 D., + (2ivFk)2 _%f2 i Yi 1 , (7)
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Since the sections of the hypothetical standard circuit are divided into ml unit
where Vi is the field attenuation factor of free space in the i - M sector.
The value 1 is the mean for one hour of accidental quantity ias-
(_1
Vhr
much as eq.(3) determines the average for one hour of noise volume.
Further,
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'where AF is the width of the line spectrum;
bmean is the equality (in nepers) between the mean power level of all
channels and the measured level of one channel*;
Y2(00, Y3(0 k) are the functional values of the distribution of spectral densi-
ty of the nonlinear products of the second and third order at a
frequency Fk of the given channel;
b21, b31 are the nonlinear fading (in nepers) at double- and triple-
frequency harmonics for stations with modulation and demodula-
tion signals (the circuit consisting of modulator, demodulator,
and group amplifier), measured for unit frequency variation Afl;
Yl, Y2 are the resolution factors (for first and second degrees of
separation) of the characteristic of the group signal time in
the channel of one section of the hypothetical standard circuit,
engaging the ml station.
Equations (7) and (8) disregard the nonlinear products formed in the antenna
? feeders as a consequence of the energy reflected from the mismatched load. If the
feeders are comparatively short, these products will mostly be nonlinear products of
the second order and may be considered as adding the following value to the right-
hand side of eq.(7):
2m, K III B fi - IF e 2 (21rF")2 (2zAf1)2
E r4 r~ti2
(9)
The Third Research Committee of the NKKF recommends that this value be considered
as equal to bmean = - 1.72 + 2 In N when there is a sufficient number of chan-
nels N. The value -1.72 neper represents the mean level (per hour of greatest
charge) of the speaking currents power in the channel at the relative zero level, de-
termined on the basis of measurements in several countries. Unfortunately, no suit-
able value has been determined for our domestic communications systems.
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where 2m1 is total number of feeders;
Tfl is the group wave transit time in the i - M feeder;
rli,r2i are the reflectance at the ends of the ith feeder.
Equation (9) is valid,.provided that
"NAf k ebmt T fl G 0,5.
Otherwise the calculation of the nonlinear products, originating in the feeders,
in eq.(3) would become extraordinarily complicated.
It is apparent from egs.(2), (5), (7), and (8) that the values x, Ao, B1, Cl
are determined by the basic parameters of the equipment, whereas the value Al - A
= AoSl lu, depends also on the radio-wave propagation conditions. Therefore, the
problem which we presented at the beginning of this article may be defined as fol-
lows: It is required to determine the values Ao, B1, C1, and x from the condition
that the noise volume in the channel at the end of the hypothetical standard circuit
is equal to the allowed value Pna = 7500 ?w?
? For one section of the hypothetical standard circuit, this condition corresponds
to the equation
where is the number of sections in the hypothetical circuit (v = 6 at N = 12 _v 60
and 'v = 9 at N > 60).
It is obvious that eq.(11) alone is not adequate for determining the four un-
knowns Ao, B12 C1, and x. We will, therefore, present a second condition, guaran-
teeing a minimum of technical difficulty in the practical construction of the equip- STAT
P not 1
-? = Bnd, = A0S1 hr x -~ B1x C~x2,
(10)
ment. For this, we assume that the value x, determined by the frequency variation,
is the optimum value and that the noise volume will thus be minimal. Differentiating
eq.(11) by x and adjusting the result to zero, we obtain the second equation
0- -/10 1 hr z? + Bi + 2Cix.
(12)
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The two equations (11) and (12) are still insufficient for solving the problem;
~ - however they will yield
B, x = 2Pnd I- - 3A?S, hr z-
CLx2 = 2Ao.Si hr x End,
which implies
Pnn -. = 2Pnd, -- Vin
Pnn, = 2Pin -- Pnd,
(13a)
Apparently, in the real system, the power of nonlinear noise of the second or-
der, as well as the power of nonlinear noise of the third order, cannot be equal to
zero and still be less than zero. Bearing this in mind, we obtain from eq.(13a) the
third condition
2 Pnd, < Pin < 3 Pnd,
? We further assume that the thermal noise volume equals
P f n _ Aosi hr x = Tnd
where the value E, in conformity with eq.(14), lies within the limits of
Then from egs.(13) and (13a) we find the-ratios between the components of the
1 2'
noise in the telephone channel
to AOSI hr -z = EP?d.,
Pnl =B,x= 31'd, C 3
(14)
(15)
STAT
Mn P =C :_ I)i 'IX2 2P,,4, , (1 2
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'where 9 satisfies the condition (15).
The ratios (16), (17),
.0
49
ae
47
a5
a4
a3
a2
Q1
504SZvo6wimawam am 4
Fig.l
and (18) are optimum since they are derived from the con-
dition that the minimum possible noise volume in the
channel of the radio-relay system is determinable by
the parameters Ao, Si hr, B1, Cl a'id eq'ia1 to the
allowable value Pndl. _
To illustrate the ratios obtained, the noise
components are plotted in Fig.1 as a function of. the
value , in which the allowable noise level is taken
as unity.
Determination of the Basic Parameters of the Equipment
We note that, of the four values Ao, B1, C1, x, determining the basic parameters
of the equipment, only one value, namely x, can be considered known, since the fre-
quency deviation of the channel has been fixed by the specifications established at
the Eighth Plenary Meeting of the MKKR in September 1956. To insure international
contact of the radio-relay lines, the MR recommends the following effective fre-
quency va or_ values of the channel 0 fk, for systems with 24 to 600 channels
(Bibl.
Number of Telephone
Channels in the System
Effective Value of
Frequency Variation
in the channel Afk (kc)
24
35
60
50, 100, 200
120
50, 100, 200
240
200
600
200
25
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Therefore the three equations (16), (17), and (18) obtained above are entirely
adequate for determining the three unknowns A0, B1, and Cl*
From this, the following equations are obtained:
A0 = P,,d, EX
91 hr
B, = Pnd, (l z 3E)
C I - Pnd,
?
S, =
6
(19)
(2O)
(21)
In these expressions, the value a is not accurately determined; it is known
only that it must satisfy the inequation (15). In selecting the value 9, the de-
signer has an opportunity of changing the numerical ratio of the noise components,.
i.e., to increase one of the parameters (B1 or Cl) at the expense of the other, if
it is expected that nonlinearity of the second or third order will be predominant.
As can be seen from eq.(19), the parameter Ao also depends on the nonlinearity
of the value S1 h,, expressing the conditions of the crossing signal for all ml
parts of one section of a hypothetical standard circuit. It is known that-the at-
tenuation factor V1, in every part of the system, is an accidental value. Thus,
- the sum
(2E - 1)
YJ 1
is also a determinant of the volume of thermal noise and likewise is an accidental
value. In order to find the probability distribution for S1, it is necessary to
know not only the laws of distribution probability for all Vi, but also the ratio
of these accidental values, since they are not independent. This is obviously a
very complicated task.
However, the allowable noise volume is given as an average per hour. Conse-
quently, as already mentioned, we must consider the mean value of the sum S1 hr per
hour, which is also an accidental value.
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Treating the problem in this manner greatly simplifies the task, since the mean
value of the sum is equal only to the sum of the mean values of the?indi idual
components
and it is not required to know the correlations between Vi in the various.parts.
Nevertheless, to determine the probability distribution of the value S1 hr the laws
in the various sections, published in numerous papers, cannot be used for determin-
ing 1 , since they do not present the statistics on an hour basis and for a
Vi hr .
long period of time. Thus, a corresponding statistical revision of existing materi-
al and a compilation of new experimental data is needed on the probability distribu-
tion of the value _. or, directly, the values of mean thermal noise volume in
V
i hr
mentioned that the statistical probability distribution curves for the values of Vi
of distribution for the values ( V 1 in all sections mast be known. It should be
\ 2
i hr
the channels of various sections of the system.
The problem may be solved by approximation, if the probability distribution of
the value 1 for some average or even for the less efficient part of the sys-
V2
i hr
ten is known. In this case, the wanted value of the sum
where the value 1 , is so selected that the accidental value 1_1 in I
V2 STAT
h
r r
the given part may exceed the one selected in the course of a fixed but small per-
centage of the time.
The recommendation by the MKKR specifies the allowable mean noise volume in the
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channel in the course of any hour. The optimum correlations listed above eqs.(16),
(17), and (18) between the noise components and the basic parameters of the equip-
ment (19), (20), (21), whose calculation is based on these correlations, refer to
the hour of greatest load, when the nonlinear noise level is at its highest. There-
fore, the value Sl hr' essential for the computation, must be specified for the day-
-light hours, when, as experience shows, the signal fading is usually not great. At
night and in- the- early morn i-ng when the fading is usually *greater, the line load is
small, nonlinear noise is almost absent and the rise in thermal noise above the op-
timum value is completely permissible.
The supplements to the recommendation by the MKKR contains examples for typical
probability distribution curves of thermal noise in the sections of the system, and
also examples of the reckoning from which it follows that, even under adverse condi-
tions, the value is 1 V2 = 2 - 3 (3 - 5) db, on the average, in each part of
mean hr
a line of 2500 km.
These considerations are also contained in the same recommendation so that a
comparison of two suggested values will give the permissible noise volume: for an
? --hour of greatest load in the absence of fading, and for any given hour.
Actually, if fading is absent, then S1 hr = ml and eq.(11), determining the
noise level, may be written as
1"d? = P = Aom, ? -F Rix. + C1X",
d? X
where Pndo = 5000 ??w, in accordance with the recommendation.
If egs.(16), (17), and (18) are substituted here, bearing eq.(22) in mind, then
it is possible to obtain quite readily
&I h' \ _ E
I)2
VY P"d.
1 Mean)hr - - 1 -F E
pnd
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With a change of 9 in the interval (15), the value ( ( V12 I will vary from 3
Pndo 5000 mean hr
to 2, if End 7500
However, this does not infer that the value S1 hr should be chosen in all cases
to be equal to S1 hr = (2 - 3)m1. It is obvious that the value S1ahr depends on the
usable wave range and has to be determined by the principles of statistical measure-
ment by various means.
Computing the values Ao, B1, an,2 C1 according to egs.(19), (20), and (21) it is
easy to determine the basic parameters of the equipment from egs.(5), (7), (8),
and (9). Then the designer still has some freedom in choosing the values of the re-
- maining parameters.
Thus, eq.(5) will yield the value which can be called the equivalent power
4AFkkn16r2R2 ( Fk 2
Pe - n e 2a'o ),2f? \ Ah
(23 )
The designer has the choice of selecting the values of the individual parame-
mains unchanged.
- ters n, Ptr, GA, resulting from the existing technical possibilities of their reali-
zation under the condition that the value of Pe, which is determined by eq.(23), re-
Exactly in the same way, the values b21 and Y 1, b31 and y2, determining the al-
lowable nonlinearity of the route, cannot be found in a simple operation from eqs.(7)
and (8), using the computed values J31 and Cl. However, by fixing the values of b21
and b31 it is possible to find y j and y2, respectively. In the process of -drafting
the equipment further, in the basic computation of the characteristics of the indi-
vidual junctures, a change may be required in any of the selected values; however
-there must also be a corresponding change in its dependent values such that the
w
on b21 and y2 on b31, for the found values of B1 and Cl.
To simplify the calculations, it is recommended to plot the functions of Y1
values B1 and Cl remain constant.
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As examples, r`igs.2, 3, and 4 show the graphs, calculated according to the for-t
mulas given, by means of which the basic parameters of the equipment can be deter-
mined.
IN -240
Q60
Q63
_ N-127
-Q56
a60
053,
Fig-3
These graphs are calculated for two radio-relay
systems with 120 and 240 telephone channels, having
the following data:
iFk = 3.1 kc; kn = 0.75; Afk = 200 kc; to =
= 707 kc (amplitude equals 1 mc); v = 9; ml = 6; Ro =
= 43.6 km; X = 7.5 cm; bmean = 1.72 + 2 In N;
1) The system with 120 channels: N = 120; Fk =
552 kc; AF = 492 kc; y2(ak) = y3(ak) = 0.45; bmean =
= 0.67 nepers;
2) The system with 240 channels: N = 240; Fk =
= 1052 kc; AF = 992 kc ; Y2 (ak) = Y3 ((Tk) = 0.5; bmean =
= 1.02 neper.
In Fig.2, the functions of the equivalent power
Pe are shown, reflected in decibels proportional to
1 watt, dependent on the mean value of signal fading
in each section I n Figs-3 and 4, the functions of Y 1 at b21 and Y2
V2
mean hr
at b31 are shown, from which it becomes quite clear that an increase in attenuation
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since Y1 and Y2 remain practically constant. _
If the nonlinear products, arising in the feeders, increase further, then the
of the nonlinearities b21 and b31 above the determined. values is of no meainiiii,
value Y determinable from the graph in Fig-3,, will equal
The values y l and Y2 are the coefficients of the group time characteristics of
the signal transit over the track of the entire section of the system, consisting
of ml stations. To find the corresponding coefficients Y li and y21 for one station,,
it is necessary to know the combination law of these coefficients. This problem is
considered in some detail in a previous paper (Bibl.3).
This expression, derived from eqs.(7) and (9), is correct for calculating the
inequation (10).
connection.
accompanied by a-sharp rise in noise, which may result in a disruption of the
tude limiter, a drop in the power of the signal at the station output will result,
level at the receiver input of any station drops below the threshold of the ampli-
The maximum amplification of the tandem office is determined primarily by the
need for stability of the connection. If, as the result of deep fading, the signal
ing of m parts represents 5% of the time, then a nonsimultaneous deep fading in the?? -
If we assume that an interruption of this type in connection in a 'line consist-
various parts, will lead to disruptions m % of the time.
Let us assume that in -4-% of the time the fading factor V in the section of
the system must be smaller than a value Vm. Then the total attenuation in the
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16n2RU 't,
)2G'2 i/2
during (100 - m )% of the time.
Obviously then, to insure a stable connection over (100 - m of the time,
the maximum amplification of the intermediary station should be equal to that fading
1(:r2/ ,'dal.
11 nlax - ' / 2 r 2 2
Min
(24)
The value Vmin is derived from the statistical probability distribution curve
of fading in' the various sections of the system.
I
1. - Eighth Plenary Meeting of the MKKR Warsaw, 1956. Permissible Noise Volume in
a Hypothetical Standard Circuit. Recommendation, Doc. No.947
Eighth Plenary Meeting of the MKKR Warsaw, 1956. Hypothetical Standard Circuit
for Broad-Band Systems with Frequency-Division Multiplex. Recommendation.
Doc. Nos.783 and 802
3. Borodich,S.V. - Calculation of the Noise in the Channels of Radio-Relay Systems
with Frequency-Division Multiplex and Frequency Modulation. Elektrosvyaz'
No-3 (1956)
4. - Eighth Plenary Meeting of the MKKR Warsaw, 195?. Connection in the Intermedi-
ate Frequency of Radio-Relay Systems with Frequency Modulation and Frequency
Division Multiplex. Recolmendation, Doc. No=730
Article received by the Editors 5 July 1956.
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?
?
AN INVESTIGATION OF THE SELF-OSCILLATING SYSTEM IN AN OSCILLATOR WITH
A' SEMICONDUCTOR JUNCTION-TYPE TRIODE
by
S.M.Gerasimov -
Phase relations are examined in the semiconductor junction-*
type triode, and a conclusion is drawn as to phase balance in the
self-oscillator. It is shown that phase balance failure may pre-
vent the generation of self-oscillation at higher frequencies.
Systems are proposed with phase correction, pamitti-.g a subst II-:--
tial boost in the critical frequency. An analysis is given of!
oscillation characteristics and recommendations are advanced, re-
lating to the selection of the optimum system of self oscillation.
An Investigation of Phase Relations
In self-oscillators, not only the energy indexes have a decisive significance
but also the phase relationships. In the investigation of oscillators with indepen-
dent excitation and semiconductor junction-type triodes, the phase relations were not
considered (Bibl.l).
In the steadily working system of the self-oscillator both phase and amplitude
balance are attained. It is known from the theory of self-oscillation that the con-
dition for phase balance is
'f., ' w,. -1 Pa = 2-I1,
(1)
where T. is the phase shift in the triode (the phase angle of average transconduc-
tance);
'Pk is the phase shift in the feedback circuit;
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T a is the phase shift in the circuit (in the load);
n = 1, 2, 3 ..., where n is a natural number series.
The phase shift in the semiconductor triode (or tetrode) differs basically from
the phase shift in an electron tube. In the semiconductor triode (in the following,
the tetrode can be substituted everywhere too) the phase shift is determined by the
effect of the current flowing through the base and the junctions between emitter and
base and base and collector.
13 If the investigation is restricted to no-voltage and critical operation, the
low permeability of semiconductor triodes will make it permissible to neglect the
cg).lector voltage response, in first approxi.*na-
tion.
Under these conditions, the equivalent tri-
ode system must be derived from an analysis of
the currents in the system shown in Fig.l.
+ tE7,
Under the effect of the excitation voltage uo
to in the circuit, the current begs to flow A
voltage drop occurs in the pedestal resistance
,
as well as in the electric wires (this is denoted
N
N
as the resistance ro). The remainder of the vol-
tage will operate at the emitter-base junction.
Assume that, under the voltage effect at the emitter-base junction
uto=U"'Sill(Ut
the following current begins to flow in this circuit:
io, = eU Sin mt.
Then, the following current begins to flow in the base-collector circuit;
tox =I sill (Lot - ).
?d~
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The phase angle Yd, is determined by' the drift time of the minority carrier
(from the emitter-base junction to the base-collector junction) and by__the frequency;
`Pdr= wldr-
The current in the base circuit, generated by the excitation source, will be
equal to the difference in currents at the junction (Fig.l)
ro=r.o--io,. =losinwt -loxsin (wt -pd,).
In the general case,
(2)
!~ :i_ I~,K?
However, at a known error it can be permitted that Ieo = Iok = Ik if the recom-
bination of the minority carrier with the base
is neglected; besides, it is assumed that the
+;xe, u,9 slope of the curve of both currents is the same.
___ Thus, if it is allowed that Ieo = Iok = Ik,
then eq.(2) will yield
Fig.2
fdrl~
i,,- 21K sin ` `a cos 1 wt--
eo
2l.. sin ar U
io =1ti, sin + Cdr cos Wt 21K sine -'sin wt.
The equivalent system, shown in Fig.2, corresponds to the existence
currents.
The equivalent resistances are determined by the conditions
Isin p _ Uto
K
Xeo
35
(3)
(4)
of these
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From the conditions of the equivalent circuit of Fig.2 it follows that
I en
rep, _ -
:.IN sill-'dr
U,Y - )
..eu - lxsiu rdr
a
the sLa Ustica c rac dE0
The ratio Ik represents the mean transconductance for the variable component
Ueo
of the collector current Ik = Smears Ueo. At the very lowest frequencies, the vol-
tage at the junction Ueo is practically equal to the voltage of the excitation source
u0. Therefore, the value of Smean should be determined at the lowest frequencies -
Ik In Class A operation, the value Smean is equal to the steepness of
as smears = uo ?
teri QM c S = dIk when
1 h
?
Fig-3
1 U6;
41.,
Fk'= const.
On computation of the-stated Values, we
i
_i'. iii ?dr
t?
(6)
(7)
At 0 cd, < no the reactive component will
be xeo < 0; under the conditions assumed by us this implies that this has the charac-
ter of capacitance.
This case is well illustrated by the discussed vector diagram (Fig-3). The
lagging vector of the current Iok corresponds to the vector leading by the phase cur-
rent Io.
It is easy to show that the angle of'drift it cps. < 2n corresponds to the re-
active resistance of an inductive character (xeO > 0). The resistance is reo > 0
at 0 < Ydr < 2 n. The general character of change in the functions reO, xeo = T ((Pdr )
is shown in Fig.4.
? Let us determine the equivalent input capacitance C. S once, in this case,
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i Xeo = - i 1 , then from eq.(6) we obtain
w 0eo
re = /,sin Tdr
Ueo
Equation (8) may be presented as
/V tdr Sin edr Q silt ?dr
CH- dr' i en 74r
(g)
(9)
-where Q = Iktdr is the charge of the minority carriers of the current in the base of
O
The capacitive nature of the
X?c?Teo I,,
----
25
%r-
i I
I
31
2
to the fact that the value of the equiva-
lent capacitance depends on the angle of
drift (I'ig.5). The equivalent inductance
may be found from eq. (6) under 'the condi-
tion that i xeo = iw Leo or
Uro
to/g Sill tdr
It should not be forgotten that,'when
?I
It < IL 21t, the co-factor of the denomi-
Fig.4 nator is sin T dr < 0 and, consequently, in
eq.(10) 1,60 > 0 is obtained.
Turning again to the system in Fig.2, we find the correlation between the vol-
tage at the junction Ueo and the voltage of the excitation source Uo. The voltage
at the junction equals
C
0 _0ixeorQO U eIito
en ,1( 7. r., -{- i x,,,
triode input resistance is well illustrated by
eq.(9). The co-factor s dr testifies
Tdr
7 = r0 - i xenreo
r., + i xeo
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0 where
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After several transformations, the following equation is obtained:
where
Equations (il) and (12), at drift angles of 0 < q . it and irrespective of the
/a0'Ydr
active resistance reo (reo >_ ro), yield the pre-
viously known fuctions (Bibl.l):
?
It is important to establish how K and Teo
change on any variation in cpdr within the limits
of 0 to 2n. Using egs.(7), (13), and (14) we get
The values of interest to us are listed in Table 1.
The greatest interest is represented by the deduction that the angle (Peo changes
within the limits of since, as indicated in Table 1, - 1 < (4-) < + 1
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[see eq.(12)].
Table 1
Tdr
P
Q
p
Ito
h
i)
I
rr
a
U
I -
rt
ruSMU~
~,~ M
I
+ r, wA.
-roSS f,"
-- t + r.S
roan
-- n rc t;; I t r,, tiw~..n
(t t rn 1 loo s
1 _
3a
roSMUh
~~ ~n+uw
I
I t- roS ~.a?
I -E- ru S.wa.w
L' (I~ ru ~~ ~ro~
vn
I
I)
U
U
I
Typical slopes of the curves P eO = (P ((P dr) ari K _ (P ((P dr) are illustrated in
At the beginning of the investigation, it was mentioned that the current in the.
Fig.6.
base-collector circuit lags in phase behind
the current in the emitter-base circuit by
an angle of P dr =w tdr. Consequently, the
overall phase shift between the voltage of
in the collector circuit Ik equals
In principle, the angle of drift can
be found analytically. In the given case,'
angle (pdr by experimental means. To calculate P dr, the schematic depicted in,Fig.? _
can be used. In this diagram, under the condition that Ro>> Zeo, the current of the
emitter will be in phase with the source voltage; the alternating voltage-in.the- cot-
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the vertical plates with the voltage Uk = IkRn,
b) 0 then the phase shift CPdr may be determined by
the slope of the axis of an ellipse.
The results of the measurement of Pdr = (f)
N R.
lector will be exactly in antiphase with the current Ik.
If, the horizontal plates of the oscillotron, are fed with the voltage Uo and
for the triode PZB are presented in Fig.8.
Eo- +E?-
Fig.7
a) Horizontal; b) To the
oscillograph; c) Vertical
The -changes are realized in the linear am-
plification system with a collector current
Iko.= 30 ma. The resistance Ro (Fig-7) will
have a value of 1 k-ohm and the resistance Rn =
Fig.-9
150 300 450 603firq a)' Vertical; b) To the oscillograph;
Fig.8 c) Horizontal
the ellipses permitted determination of four frequencies at which the angles of
_ phase shift O dr) were equal to 0, 45, 90, 135, and 180?. An oscillator of the type
I-100 served as the oscillating source.
= 300 ohms. The driving voltages and the loads from the terminals were fed-directly
to the plates of the oscillotrons (osci]lograph DO-4). The specific arrangement of
30
PZB
(50
P' Y(f)
i
i
99
60
}
The total angle fps = ro eo + T dr can be determined from the diagram in Fig.9. In
this diagram, the overall phase shift `ps =`peo + pdr is equal to the shift in phase
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between the excitation voltage U0 and the current of the collector Ik (or the voltage
under load Un = IkRn).
This phase shift is easily determined by means of the oscillotions (see the
test for calculating Tdr), or the voltages are given directly at the vertical and
horizontal tube plates, passing through the amplifier.
The result of the variations in the angle T. for the triode PZB are given in
Fig.lO. The measurements are made by the very same method used in the previous ex-
periment. It is significant to compare the measured angles Tdr and T. at a frequency
of approx. 170 kc (Figs.B and 10). Here we have the angle Tdr = 900 and the angle
Ts = 1350, which, generally, agrees quite well with the theoretical conclusions (see
Fig.6, where Teo < 45? when qdr = 900).
At the lowest frequencies, the angle Teo is quite small, and consequently the
values of the angles Tdr and T. converge
Y
0 (see Figs.8 and 10, for f < 50 kc).
At the highest frequencies (f >
135 1
75 E- gles Pd, and Ts again converge (Figs.8
if 90 200 300 4anikc and 10)
The total phase shift Ts = Teo +
Fig.10
+ Tdr is the shift in phase between the
voltage of the excitation source UO and the first harmonic of the collector current
Ikl. Applicable to the self-oscillator layout, this phase shift is depicted in
Fig.ll.
In the known systems of vacuum-tube self-oscillators, the phase shift between
the currents Ial and the voltage under load (circuit) U usually'tends toward zero.
This indicates that the frequency of the generated oscillations and the natural
frequency of the circuit practically coincide. From. the view-point of the required
41
> 400 kc) the angle Teo is greatly re-
90 -
duced (Fig.8) and the values of the an-
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frequency stability and energy indexes, this system is optimal. However, it is
known that, if the sum of the phase angles of the triode Ts and the feedback Tk do
_ ? not equal zero, a separation between the natural and oscillating frequencies will
occur. The sign and value of the separation are determined by the equation
(20)
where q'a is the load phase angle.
In principle, the angle ma may vary within the limits of t90?; however, in prac-
tice by virtue of a considerable reduction
of the equivalent resistance of the cir-
cuit outside of the limits of its zone of
passage, the angle q'a changes within con-
siderably smaller limits. In other words,
if it is required to satisfy the phase
balance at greater T a, then the oscilla-
Fig.11
-0
tion collapses due to a disrupted amplitude balance.
It follows from this that the sum of the angles q)s and T k must also be close to
zero [see eq.(20)]. The variations in the angle cps for the semiconductor triode
were already examined.
In the known systems of single-circuit self-oscillators, the phase angle of the
feedback factor can be made to approach 00 or 180?. In two-circuit and three-circuit
self-oscillators, the phase of the feedback factor, at loose coupling between the
circuits, may vary within wider limits due to a variation in natural frequency of
one of the circuits (in the two-circuit system) or of two of the circuits (in the
three-circuit setup).
From the above statement it follows that the values (ps + (Pk approaching 90? are
more disadvantageous, since it is hardly possible to compensate for such a shift in
e
phase [sqe eq.(20)] in the common self-oscillator. Consequently, in self-oscillators
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18
with a semiconductor triode, one may expect oscillation failure at frequencies to
which q)s + Tk - 900 corresponds.
As mentioned above, in ordinary self-oscillation systems, the value CPk - 0 can
be considered as the first approximation. Then, the approach of P s to 900 denotes
the limiting frequency of self-oscillation, if the characteristic cp s - p(f) is used.'
Obviously, an oscillation failure may oc-
cur due to an amplitude-imbalance. How-
ever, we will disregard this point for the;
time being.
To put all the above material to a
practical test, a self-oscillator was se-
lected with an inductive Hartley circuit
(Fig.12) in a triode of the PZB type. For
this, the earlier characteristic s = P (f )
Fig.12
?
(Fig.10) was taken. -
Careful measuring showed that the highest cutoff frequency of the oscillator fnp
is 90 kc. A study of the curve (Ps = (P(f) shows that the angle P s = 900, is corre-
lated by a frequency of, for example, 70 kc. The coincidence of the results should
be considered satisfactory, if the error in measuring the phase P s is taken into con-
sideration according to our accepted method (by placing the large elliptical axis at
the tube grid). Besides this, the capacitance Co of the separating capacitor (see
below) could affect the result.
Investigation of Phase Correction Systems
For phase relation changes in the self-oscillation system, some reactance should
be included in the base circuit (Fig.13).
If the following designates
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then it is not difficult to obtain the following expression for the phase angle:
r r -- xxe?,__
arctt; -- -,
rC'?xu I. 1e? ra -I- 'Ye , r
where ro, reOf xo, xeo, see Fig.13.
(21)
Let us examine the variation limits of the angle 9eo in the case where 0 : Tdr<
< n. In this case, we have xeo < 0 (of a capacitive nature). The nature of the val-
ue changes xeo - T(Pdr), reo = P(9dr) is
+jXn 7o shown in Fig-4.
fi Let there be some auxiliary capaci-
00 lea 4iXeg
U
----__~en tance included in the base circuit, i.e.,
i let x0 < 0. Then, the denominator in
Fig.13
eq.(21) will always be less than zero. The
subtrahend in the numerator of eq.(21) will be greater than zero (xoxeo > 0). There-
fore, at
rure? - ' x,. Xe? > 0
(22)
the phase angle T? will be optimal, i.e.,
Ueo will lag in phase behind U0; and at
the phase angle Teo will be positive, i.e.,
Ueo will lead Uo in phase. The first case
Fig. 14
Actually, in this case,
creased, inasmuch
xo
. r(r? --- xvxe? < 0
corresponds to the large capacitance Co
(Fig.12), and the second to the small ca-
pacitance. The best results are obtained
with a low capacitance Co.
will be large, and xeo will also be greatly in-
as, at the higher frequencies, we have P dr it (see Fig-4). The
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value reO, on the other hand, will tend toward the minimum.
Therefore, the fractional numerator in eqs.(21) will increase rapidly, while
the denominator will rise more slowly. The absolute value of the fraction will in-
crease with decreasing Co, and a boost in frequency (increase in Tdr), i.e., an ad-
vance in Ueo relative to Uo, will also increase the value. In Fig.11 the possibili-
ty of maintaining phase balance (q)s < 90?), even at angles (Pdr approaching-1800, is
demonstrated. As a consequence, a considerably more favorable phase balance can be
expected in the system with a low capacitance Co. As a result, higher frequencies
can be generated.
Actually, with a capacitor of Co = 1000 NNf at the triode PZB, steady self-
oscillations took place, going as low as a frequency of 300 kc. This limit frequency
trebled the cutoff frequency of 90 kc, obtained with the separation capacitor Co =
= 5? f when the condition of phase correction was not satisfied.
It would be quite interesting to analyze the experimental data given in Table 2.
This Table lists the self-oscillating frequencies and the smallest corresponding
feedback factors, insuring operation of the system close to oscillation decay. Be-
sides this, the Table gives the values of the capacitance Co, supplying the required
phase correction. The values of Co are not critical. A reduction in Co is a condi-
tion for raising the cutoff frequency of self-oscillations.
h.nr - hum
CO' pet
rl?n
110
0,002.-) 10 000
205
(),113 1 10 000
23?1
(1,0011; 5 000
2M8
(1, 015 1 000
30t 1
1()11()
a) Frequency of Self-Oscillation, in kc
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A frequency of 205 kc should be considered a limit in this respect for Co =
= 10,000 ??f; a frequency of 238 kc, for Co = 5000 ??f; and a frequency of 300 kc,
for Co = 1000 if. As preliminary data, a further decrease in Co does not permit an
-. increase in the cutoff frequency of self-oscillation.
The voltages Uom and Um are measured by vacuum-tube voltmeters VKS-7b, installed
as shown in Fig.12. Lower values of the feedback factor, at Co = 5 ?f and Co = 1?f,
-and also the corresponding cutoff frequencies are shown in Table 3.
Table 3
Triode PZB
tih (L 17 ~
7h 11, llbh
`111 Il, 1.1
51i ~1,1111'.i
77, 11 Il;i.i
cl; Il I,;
C)
a) Frequency of self-oscillation, in kc; b) Remarks; c) Cutoff frequency
In addition to its usefulness as a phase correction circuit, working with a low-
capacitance capacitor in the base circuit Co, the phase correction system can be op-
erated by means of inductance, and, consequently, a large capacitance Co can be con-
tained in the capacitor. Equation (21), derived above for (Peo, indicates that in
this case (and even at 0 . Tdr n) the products xoxeO < 0 and the fractional numera-
--tor (21) can be converted to zero, if the change x0 = WL0 (assuming C - 0) re-
W0
sults in the conditions roreo - xoxeo = 0. The fractional denominator (21) may be
equated to zero when reo xo = xeo r0 + xeo reo are larger or less than zero. At
? -large enough a value of x0 the numerator of the fraction (21) will be smaller than
zero, and the denominator larger than zero. Thus the fraction will be smaller than
? zero and the phase anglecpeo will be less than zero. Ueo will lag in phase behind U0.
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0
The corresponding vector diagram is shown in Fig.15. From this may be inferred
that, in the large zone of the angles Pf,, phase balance may be obtained in the self-
Fig.15
I
oscillator, constructed as an "inverted" circuit:
Such a hookup (Fig.16) was tested and showed
good results. In the absence of resonance
(roreo - xoxeo Y 0), with inductance in the base
circuit Ib, a frequency of self oscillations of
320 kc (with a PZB triode) was reached. In the
presence of "resonance" (roreo - xox, = 0), a
frequency of 500 kc was obtained. The required
information is presented in Table 4.
Table 4
Triode PZB
?
A)
Uuttt
?uC = Um
Lo, y henry
(()
0,02
100
82
0,007
100
C)
105
0,0033
100
150
0,0053
100
230
0,012
100
:320
0,03
100
500 ----
0,062
e)
a) Frequency of self-oscillation, in kc; b) Remarks; c) Lowest value of feedback
coefficient; d) "resonance"; e) Limit frequency
At the conclusion of the experiment with the system in Fig.16, the triode P1V,
was tested with which self-oscillations of a frequency of 1.4 me were obtained. At
this, the inductance, corresponding to the "resonance", was equal to 2 microhenry.
. Oscillation Characteristics
In installing a self-oscillating system it is essential to satisfy the require-
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ments not only as to phase balance but also as to amplitude. General directives in
this relation can be obtained from the previously diac,.ssed oscillatory characteris-
tics Ikl = cp(Uom). Since the voltage at the terminals of the collective load (cir-
cuit) is proportional only to the harmonic (Uk = Ikl Re), the oscillatory character-
Fig. 16
istics were assumed as Uk = T(UOM) for Re,,
Ek, Bo = const.
The characteristics for the triode PZB
were also taken at a frequency of 23.4?kc,
for the different voltage displacements at
the r>se (Eo = 0; +0.2; -0.2 v, cf.
Fig. 17).
Attention should be paid to the fact
that soft self-excitation is possible only when there is a'small emission with nega-
tive bias in the base. This result could have been foreseen, since the characteris-
tics Ik = cp(Eo), at Ek = const start
= 0 and are distributed in the
at E
0
i f -23,4 k e Ea--QZv E, ,-O
E,.-+O?v
negatively charged area in the base.
Greater steepness of the characteristic
4)/,;
S = - (~~? . , necessary for self-
excitation, corresponds to a negative
value of Eo.
For this reason, the systems of
Q5 1,0 1,5 _2.0 U (v)
Fig.17 self-oscillation presented above always
contain a preliminary negative bias at
the base. This is naturally fed from the common source Ek.
Conclusions
-1. In self-oscillators with semiconductor junction-type triodes, a disruption
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?
in phase balance may be the cause of oscillation decay at high supercritical fre-
quencies, even leading to an upsetting of the amplitude balance as a result of the
power drop of the amplification factor.
2. The use of self-oscillator systems with phase correction in the feedback cir-
cuit permits an increase by several times in the cutoff frequencies=of self-'
oscillation. Fop example, in the self-oscillator with a low-frequency triode PZB,
steady oscillations at frequencies up to 500 kc can be obtained, whereas without cor-
rection, the cutoff frequency of self-oscillation equals 90 kc.
3. To improve self-excitation conditions in receiving soft self-excitations, it
is necessary to feed the base of the triode with some small preliminary negative
bias.
?
1. Gerasimov,S.M. - Energy Indexes of Semiconductor Oscillators with Independent
Excitation in the Supercritical Frequency Range. Elektrosvyazt, No.9 (1956)
Article received by the Editors, 13eJuly 1956.
0
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I?
FREQUENCY BAND OF RADIOTELEGRAPH SIGNAL TRANSMISSIONS WITH AMPLITUDE
AND FREQUENCY KEYING
by
M.S.Gurevich
C
The frequency bandwidth is defined which is assigned to
radiotelegraph transmissions, with amplitude and frequency keying
by rectangular and rounding-off signals.
Introduction
The frequency bandwidth of transmission is important in characterizing the sys-
tem of radio communication and in indicating the effectiveness of the use of the
frequency spectrum. The bandwidth assigned to transmissions is called the frequency
band, containing 99% of the transmission power and including any discrete frequency
whose power consists of not less than 0.25 of the common transmitted output (Bibl.l
and 2).
. The examination presented below is based on the spectra of signal transmissions
("telegraph points"). This method, corresponding to the requirement for "regulation
of radio communications" (Bibl.l), easily permits theoretical computations and cor-
responding measurements. It is known that the spectra of "telegraph points" and the
actual communications are closely adjacent. The establishing of closer relations
between assigned bandwidth with the transmissions of "telegraph points" and the ac-
tual communications demands a special study and is beyond the scope of the present
article.
Bandwidths Assigned to Transmissions with Amplitude Keying
r It is known that the distribution of energy in amplitude-keyed oscillations is
the-very'same as in the low-frequency keyed signal, excluding the constant multi-
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f(t)
(ple 1/2 which is kept in relation to the keyed oscillations. In the following, the
spectra of low frequency signals will be. considered as their equivalent..__.___. ,_-I
?
will yield
Fig.l
Square-rave sisals. When keying by
!square-wave signals (Fig.!)
RO 1o
at 0